Current converting method, transconductance amplifier and filter circuit using the same

ABSTRACT

The present invention is intended to achieve a transconductance amplifier and a voltage/current converting method which can provide a sufficient amplitude and a high degree of design freedom. The method comprises the steps of converting a first voltage signal to a first current signal; converting a second voltage signal to a second current signal; obtaining the common-mode components of the first and second current signals; and subtracting the common-mode components from the first and second current signals to obtain third and fourth signals, and further, subtracting the fourth current signal from the third current signal to generate a first output, while subtracting the third current signal from the fourth current signal to generate a second output.

TECHNICAL FIELD

The present invention relates to a transconductance amplifier and avoltage-to-current converting method which are effective in RF low-passfilter circuits.

BACKGROUND ART

In recent years, RF systems have been used as means for communicatinginformation between devices in portable electronic devices, electrichome appliances, peripheral devices for personal computers, and thelike. Wireless systems used in these electronic devices are fabricatedin semiconductor integrated circuits to reduce size and weight and tolower price. Generally, an RF system requires a filter which has anabrupt cut-off frequency in order to separate a particular frequencycomponent. However, since elements used in semiconductor integratedcircuits suffer from substantial variability in manufacturing, it hasbeen difficult to accomplish a filter circuit which has an abruptcut-off frequency. Accordingly, a Gm-C filter composed of atransconductance amplifier [hereinafter called “OTA” (OperationalTransconductance Amplifier) and a capacitance has been used.]

FIG. 1 is a diagram showing the basic configuration of atransconductance amplifier (hereinafter called “OTA” (OperationalTransconductance Amplifier).

As illustrated, the OTA is an element for generating currentsG_(m)V_(in)/2, −G_(m)V_(in)/2 which are proportional to input voltageV_(in), and ideally has infinite input impedance and output impedance.

In this event, proportion coefficient G_(m) is a parameter called mutualconductance, and an OTA applied to a filter and the like is configuredto have the ability to control the mutual conductance with a signal fromthe outside.

FIG. 2 is a circuit diagram showing a specific configuration of an OTAwhich is controlled mutual conductance G_(m), for example, a degenerateddifferential OTA presented in Bran Nauta, “Analog CMOS Filters for VeryHigh Frequencies”, Kluwer Academic Publishers, 1993, pp. 87-88.

Current sources 404, 405, 406, 407 each apply the same current value.Also, a resistive component of variable resistive element 403 connectedto sources of input transistors 401, 402 varies in resistance inresponse to mutual conductance control signal 408 applied from theoutside.

When input transistors 401, 402 have large transconductance, a currentof ΔV/R/2 appears at the output, where R represents a resistivecomponent of variable resistive element 403. Here, ΔV represents avoltage of a differential component of a voltage signal applied to theinput. Therefore, arbitrary mutual conductance G_(m) can be accomplishedby controlling the resistance of variable resistive element 403 withcontrol signal 408. This configuration of an input stage is generallycalled a total differential input stage.

With the miniaturization of processes in recent years, lower voltageshave been required for power supplies. In particular, when a sourcevoltage is equal to or lower than one volt, limitations are imposed onthe number of vertically stacked stages of transistors which can be usedbetween the power supply and GND, thus making it impossible to useconventional circuit configurations. In the circuit configuration shownin FIG. 2, at least one or more transistors are required in the currentsource, three stages of transistors are used between the power supplyand GND when they are vertically stacked. A transistor generallyrequires a drain-source voltage of approximately 200 mV when it operatesin a saturation region, so that when transistors are vertically stackedin three stages, 600 mV is required irrespective of the magnitude of thesignal. Accordingly, when the source voltage is one volt, only 400 mV isavailable for a signal amplitude, giving rise to a problem of theinability to provide a sufficient amplitude.

FIG. 3 includes diagrams showing the configuration of apseudo-differential input type OTA disclosed in Ahmed Nader Mohieldin,“Nonlinear Effects in Pseudo Differential OTAs with CMFB”, IEEETransactions on Circuits and Systems, Vol. 50, No. 10, October 2003, pp.762-769, where FIG. 3 a is a circuit diagram, and FIG. 3 b is anequivalent circuit diagram of an output stage.

In FIG. 3 a, sources of p-moss transistors M_(2A), M_(2B), M_(02A),M_(02B) are commonly connected to a power supply, while source of n-mosstransistors M_(1A), M_(1B), M_(01A), M_(01B) are commonly grounded. Eachdrain of p-mos transistor M_(2A), M_(2B), M_(02A), M_(02B) is connectedto each drain of n-moss transistors M_(1A), M_(1B), M_(01A), M_(01B),while gates of p-moss transistors M_(2A), M_(02A) are connected to adrain of p-moss transistor M_(02A), and gates of p-mos transistorsM_(2B), M_(02B) are connected to a drain of p-mos transistor M_(02B), toform a current mirror circuit. Each gate of n-moss transistor M_(1A),M_(01A) is connected in common, and each gate of n-mos transistorsM_(1B), M_(01B) is connected in common to define an input part of gatesignals V_(a), V_(b). Output current Iout1 is generated from the drainsof transistors M_(1A), M_(2A), while output current Iout2 is generatedfrom the drains of transistors M_(1B), M_(2B).

In the circuit configured as described above, transistors M_(1A), M_(1B)are input transistors, transistors M_(01A), M_(01B) convert common-modecomponents of a signal from a voltage to a current, and a currentproportional to the common-mode components is supplied to transistorsM_(1A), M_(1B) by the current mirror circuit composed of transistorsM_(02A), M_(02B) and transistors M_(2A), M_(2B).

A current flowing through a current source has a current value one halfof a current value proportional to the common-mode components of thesignal generated by transistors M_(01A), M_(01B).

Generally, drain current ID of a MOSFET is represented by:

ID=½μC _(ax) [W/L](V _(GS) −V _(T))²

When the drain current of a transistor applied with a gate-sourcevoltage V_(a) is I₁, and the drain current of a transistor applied witha gate-source voltage V_(a) is I₂, where the size (W/L) of eachtransistor is the same, I₁, I₂ can be represented in the followingmanner through simplification of the above equation:

I ₁ =k(V _(a) −V _(T))²

I ₂ =k(V _(b) −V _(T))²

Here, when V_(c)=V_(a)+V_(b),

I ₁ −I ₂ =k(V _(c)−2V _(T))(V _(a) −V _(b)),

and difference ΔI of the current is represented by:

ΔI=G _(m)(V _(a) −V _(b))

As shown in this equation, difference ΔI of currents which flow throughthe two types of transistors has a value proportional to the differencebetween gate signals V_(a), V_(b) applied to their gates, thuspermitting the transistors to act as an OTA.

The example shown in FIG. 3 will be described. Assuming that transistorsM_(1A), M_(1B), M_(01A), M_(01B) are equal in size to one another, andM_(2A), M_(2B), M_(02A), M_(02B) are equal in size to one another, draincurrent I₁ flows into transistors M_(1A), M_(01A) applied with signalV_(a) between the gate and source, while drain current I₂ flows intotransistors M_(1B), M_(01B) applied with signal V_(b) between the gateand source, respectively. Since drain currents I₁, I₂ of M_(01A),M_(01B) are mirrored by the current mirror circuit composed of M_(2A),M_(02A) and M_(2B), M_(02B), a current of (I₁+I₂)/2 flows into thedrains of transistors M_(2A) and M_(2B), respectively. Here, since thedrain currents of transistors M_(1A) and M_(1B) are I₁, I₂,respectively, currents of (I₁−I₂)/2, (I₁−I₂)/2 are output from theoutput stage of FIG. 3 b. Thus, since the difference between I₁ and I₂is output in proportion to the difference between gate signals V_(a),V_(b), they act as an OTA. In this regard, in the exemplary OTA shown inFIG. 3, when the mutual conductance is changed from the outside, it canbe controlled by controlling an common-mode bias voltage of an inputsignal.

FIG. 4 is a diagram showing a pseudo differential input type OTA shownin FIG. 3 in a functional block form. In FIG. 4, first voltage/currentconverting element 1701 and third voltage/current converting element1703 correspond to transistors M_(1A), M_(1B), while secondvoltage/current converting element 1702 and fourth voltage/currentconverting element 1704, which form part of common-mode currentgenerating part 1705, correspond to transistors M_(01A), M₀₁. Currentmirror circuit 1706, which corresponds to transistors M_(02A), M_(02B)and transistors M_(2A), M_(2B), is a circuit for inverting the polarityof an input current to output a current which is proportional to theinput current.

As described above, the output impedance of the OTA is ideally infinite.As such, in the circuits shown in FIGS. 1 to 3, an output DC biasovershoots to the power supply side or to the ground side, beingincapable of acquiring a signal. Accordingly, a CMFB (Common Mode FeedBack) circuit is known for setting an output DC bias (see Non-PatentDocument 2).

FIG. 5 includes diagrams showing the configuration of a CMFB circuit,where FIG. 5 a is a block diagram conceptually showing the configurationof the CMFB circuit, FIG. 5 b is a circuit diagram showing a specificconfiguration, and FIG. 5 c is a block diagram showing an exemplaryapplication of the CMFB circuit.

First, the operation of the CMFB circuit will be described withreference to FIG. 5 a. Common-mode bias detecting circuit 703, whichforms part of CMFB circuit 702, is applied with outputs V_(OUT)+,V_(OUT)− of OTA 701, and feeds an common-mode bias component of themback to OTA 701 as output bias control signal 704. In addition to outputbias control signal 704, OTA 701 is applied with reference signal 705 asa control signal, such that OTA 701 compares output bias control signal704 with reference signal 705 to control its output such that outputbias control signal 704 provides a predetermined constant bias.

In this regard, while the CMFB circuit can refer to an common-mode biasdetecting circuit provided externally to the OTA, as shown in FIG. 5 a,the CMFB circuit includes circuits within the OTA for receiving theoutput bias control signal and reference signal for performing thecomparison and feedback, in addition to the common-mode bias detectingcircuit.

As shown in FIG. 5 b, this conventional example comprises n-mostransistors M_(3A)′, M_(3A), M_(03A), M_(2A), M_(3B)′, M_(3B), M_(03B),M_(2B) and p-mos transistors M_(04A)′, M_(4A), M_(04A), M_(1A),M_(04B)′, M_(4B), M_(04B), M_(1B).

Respective p-mos transistors M_(1A), M_(1B), M_(04A), M_(04B), M_(4A),M_(4B) correspond to n-mos transistor M_(2A), M_(2B), M_(03A), M_(03B),M_(3A), M_(3B), and these corresponding transistors have a common drain,and are provided between the power supply and ground to form an OTA.P-mos transistors M_(1A), M_(1B) and n-mos transistors M_(2A), M_(2B)make up an input differential pair, and V_(IN)+, V_(IN)− are supplied tothe gates of p-mos transistors M_(1A), M_(1B). Transistors M_(1A),M_(2A) have their drains connected to the gates of transistors M_(2A),M_(03A), M_(3A), while transistors M_(1B), M_(2B) have their drainsconnected to the gates of transistors M_(2B), M_(03B), M_(3B).

Each gate of p-mos transistors M_(1A), M_(1B), M_(04A), M_(04B), M_(4A),M_(4B) is made common, serve as node V_(X) (preceding stage), and isconnected to the drains of transistors M_(04A), M_(04B). TransistorsM_(3A), M_(3B), M_(4A), M_(4B) make up an output stage of the OTA, wherethe drains of transistors M_(3A), M_(3B) are used for an output node ofV_(OUT)+, while the drains of transistors M_(4A), M_(4B) are used for anoutput node of V_(OUT)−.

Transistors M_(3A)′, M_(3B)′, M_(4A)′, M_(4B)′ form part of the CMFBcircuit, where transistor M_(3A)′ which is supplied with referencesignal V_(Y) at a gate has a source grounded, and has a drain connectedto the drains of transistors M_(3A), M_(4A). Transistor M_(3B)′ which isapplied with reference signal V_(Y) at a gate has a source grounded, andhas a drain connected to the drains of transistors M_(3B), M_(4B).Transistor M_(4A)′, the gate of which serves as node V_(X) (next stage),has a source connected to a power supply, and a drain connected todrains of transistors M_(3A), M_(4A). Transistor M_(4B)′, whose gateserves as node V_(X) (next stage), has a source connected to the powersupply, and a drain connected to drains of transistors M_(3B), M_(4B).

In FIG. 5 b, a circuit made up of transistors M_(1A), M_(1B), M_(2A),M_(2B) is a circuit at an input stage for generating V_(a), V_(b) at thegates of transistors M_(2A), M_(2B), and is a circuit corresponding to acircuit for generating V_(a), V_(b) in the circuit diagram shown in FIG.3 a. Other parts corresponding to the circuit diagram of FIG. 3 a are asfollows.

Transistors M_(03A), M_(03B), M_(04A), M_(04B) in FIG. 5 b correspond totransistors M₀₁, M₀₂ in FIG. 3 a, and transistors M_(3A), M_(3B),M_(4A), M_(4B) correspond to transistors M₁, M₂ in FIG. 3 a. Also, amongthe transistors shown in FIG. 5 b, transistors M_(3A)′, M_(3B)′,M_(4A)′, M_(4B)′ which have no corresponding transistors in FIG. 3 aform part of the CMFB circuit.

Next, the operation of the circuit shown in FIG. 5 b will be described.

The operation of the OTA part in this conventional example is similar tothe operation described with reference to FIG. 3 a. As V_(IN)+, V_(IN)−are applied to input transistor pair M_(1A), M_(1B), V_(a), V_(b) aregenerated at the gates of transistors M_(03A), M_(03B), and areconverted from voltage to current to remove a differential component atnode V_(X) (preceding stage). When OTAs are connected in series at twoor more stages, for example, OTA1 and OTA2 are connected in series attwo stages as shown in FIG. 5 c, V_(X) (preceding stage) of OTA2 isconnected to node V_(X) (next stage) of OTA1 which is provided at thepreceding stage. In this conventional example, an common-mode biascomponent of the output signal of OTA1 appears at node V_(X) (precedingstage) of OTA2. By returning this common-mode bias component to nodeV_(X) (next stage) of OTA1, a negative feedback is applied to an outputcommon-mode bias of OTA1. Also, in this event, by supplying referencesignal V_(Y) to the gates of transistors M_(3A)′, M_(3B)′, thecommon-mode biases of outputs V_(OUT)+, V_(OUT)− are set atpredetermined biases.

Non-Patent Document 1: Bran Nauta, “Analog CMOS Filters for Very HighFrequencies”, Kluwer Academic Publishers, 1993, pp. 87-88

Non-Patent Document 2: Ahmed Nader Mohieldin, “Nonlinear Effects inPseudo Differential OTAs with CMFB”, IEEE Transactions on Circuits andSystems, Vol. 50, No. 10, October 2003, pp. 762-769

DISCLOSURE OF THE INVENTION Problems to be Solved by the Invention

In the related art shown in FIG. 2, even if the common-mode biascomponent of an input signal changes, the common-mode component of acurrent flowing through a transistor is constant with the aid of acurrent source connected to the source of the input transistor, butthere is a problem that a sufficient amplitude cannot be providedbecause a high source voltage is required.

In the related art shown in FIG. 3, the operation can be performed at alow source voltage, but since there is no current source at the sourceof a transistor which is at the input stage, the common-mode componentof a current flowing through the transistor changes depending on thecommon-mode component of the input signal. For this reason, there is aproblem that its changing amount appears at the output of the OTA as ancommon-mode signal. When the common-mode component of the signal appearsat the output, fluctuations in the operating point of the signal cancause a reduced signal dynamic range, an error in a differential signal,and even oscillations in the worst case, so that it is deemed asdesirable that the gain be at least one tenth or less. While a generalOTA-based circuit is designed to remove common-mode components ofsignals as much as possible, in the related art shown in FIG. 3, it is,however, difficult to sufficiently remove the common-mode component.

In the OTA which is provided with a CMFB circuit for setting a DC biasof the output, shown in FIG. 5, the drains of transistors M_(3A),M_(3B), M_(4A), M_(4B), which serve as the output node of the OTA, areprovided with transistors M_(4A)′, M_(4B)′ which supply a signalindicative of an common-mode bias component from the OTA provided at thenext stage when connected in series, and are provided with transistorsM_(3A)′, M_(3B)′ which supply a reference signal. Thus, four transistorsare connected to the output node, resulting in a parallel connection ofthe output conductance and parasitic capacitance of each transistor.This leads to a problem that the output impedance of the OTA is reducedsuch that it degrades the characteristics as the OTA.

The present invention has been made in view of the problems inherent tothe related art as described above, and has an object to accomplish atransconductance amplifier and a voltage-to-current converting methodwhich are capable of reducing the common-mode component of a signalappearing at an output to provide a sufficient amplitude.

Also, the present invention has been made in view of the problemsinherent to the related art as described above, and has an object toaccomplish an OTA which comprises a CMFB circuit and is prevented fromdegrading the characteristics as an OTA.

Means for Solving the Problems

A voltage-to-current converting method of the present invention is avoltage-to-current converting method for generating a first current anda second current proportional to the difference between an input firstvoltage signal and second voltage signal, characterized by comprisingthe steps of:

converting the first voltage signal to a first current signal;

converting the second voltage signal to a second current signal;

generating an common-mode component of the first current signal and thesecond current signal;

generating a third current signal and a fourth current signal bysubtracting the common-mode component from the first current signal andthe second current signal, respectively, generating a first output bysubtracting the fourth current signal from the third current signal, andgenerating a second output by subtracting the third current signal fromthe fourth current signal.

A transconductance amplifier of the present invention is characterizedby comprising:

a first and a second voltage/current converting element for converting afirst voltage signal to a current signal;

a third and a fourth voltage/current converting element for converting asecond voltage signal to a current signal;

an common-mode current generating part for converting each of the firstvoltage signal and the second voltage signal to a current signal, andfurther generating an common-mode current in accordance with ancommon-mode component of each current signal;

a first current circuit for subtracting the common-mode component by thecommon-mode component generating part from each current signal convertedby each of the first to fourth voltage/current converting elements;

a second current circuit for supplying the difference between a currentsignal by the first voltage/current converting element from which thecommon-mode component is subtracted by the first current circuit and acurrent signal by the third voltage/current converting element as afirst current output; and

a third current circuit for supplying the difference between a currentsignal by the fourth voltage/current converting element from which thecommon-mode component is subtracted by the first current circuit and acurrent signal by the second voltage/current converting element as asecond current output.

In this event, the common-mode current generating part may comprise afifth voltage/current converting element and a sixth voltage/currentconverting element for converting the first voltage signal and thesecond voltage signal to current signals, respectively.

Further, the first to sixth voltage/current converting elements maycomprise a first to a sixth first conductivity type transistor suppliedwith the first voltage signal or the second voltage signal at bases orgates,

the first current circuit may comprise a plurality of secondconductivity type transistors, the plurality of second conductivity typetransistors may have their gates in common, and at least one of theplurality of second conductivity type transistors may have a gateshort-circuited to a drain, and

an output of the second conductivity type transistor may be connected toany of the outputs of the first to sixth voltage/current convertingelements.

Also, the first current circuit may comprise a first to a sixth secondconductivity type transistor provided between a power supply and aground together with the first to sixth voltage/current convertingelements,

the second conductivity type transistors may have their gates andsources in common, and at least one of the second conductivity typetransistors has a gate short-circuited to a drain, and

outputs of the second conductivity type transistor may be connected tooutputs of the first to sixth voltage/current converting elements,respectively.

Also, the fifth first conductivity type transistor and the sixth firstconductivity type transistor may be first transistors which have thesame size with each other,

the first to fourth first conductivity type transistors may be secondtransistors which have the same size with one another,

the fifth second conductivity type transistor and the sixth secondconductivity type transistor may be third transistors which have thesame size with each other,

the first to fourth second conductivity type transistors may be fourthtransistors which have the same size with one another, and

the ratio of the size of the first transistor to the size of the secondtransistor may be equal to the ratio of the size of the third transistorto the size of the fourth transistor.

Also, the first first conductivity type transistor may form part of afirst current output supplying part, and the fourth first conductivitytype transistor forms part of a second current output supplying part,

the second current circuit may comprise a seventh first conductivitytype transistor having an output in common with an output of the firstfirst conductivity type transistor, and an eighth first conductivitytype transistor having an output and a gate in common with an output ofthe third first conductivity type transistor and a gate of the seventhfirst conductivity type transistor,

the third current circuit may comprise a ninth first conductivity typetransistor having an output in common with an output of the second firstconductivity type transistor, and a tenth first conductivity typetransistor having an output and a gate in common with an output of thefourth first conductivity type transistor and a gate of the ninth firstconductivity type transistor,

the second, third, fifth, and sixth first conductivity type transistorsmay be first transistors which have the same size with one another,

the first first conductivity type transistor and the fourth firstconductivity type transistor may be second transistors which have thesame size with each other,

the eighth first conductivity type transistor and the tenth firstconductivity type transistor may be third transistors which have thesame size with each other,

the seventh first conductivity type transistor and the ninth firstconductivity type transistor may be fourth transistors which have thesame size with each other,

the fifth second conductivity type transistor, the sixth secondconductivity type transistor, the second second conductivity typetransistor, and the third second conductivity type transistor may befifth transistors which have the same size with one another,

the first second conductivity type transistor and the fourth secondconductivity type transistor may be sixth transistors which match insize with each other, and

the ratio of the size of the first transistor to the size of the secondtransistor, the ratio of the size of the third transistor to the size ofthe fourth transistor, and the ratio of the size of the fifth transistorto the size of the sixth transistor may be equal.

Also, a transconductance amplifier may comprise a plurality oftransconductance amplifiers described in any of the foregoing,

a fourth current circuit provided in one transconductance amplifier forgenerating a difference between a first current output and a secondcurrent output in the one transconductance amplifier as a first currentoutput; and

a fifth current circuit provided in another transconductance amplifierfor generating a difference between a second current output and a firstcurrent output in the other transconductance amplifier as a secondcurrent output.

Further, the common-mode current generating part may comprise a seventhvoltage/current converting element supplied with a third voltage signalat a base or a gate, and generate an common-mode current including abias current in accordance with the third voltage signal as thecommon-mode current.

Also, the transconductance amplifier may comprise a first bias currentgenerating element for generating a first bias current added to areference current supplied by the first current circuit to the secondcurrent circuit; and

a second bias current generating element for generating a second biascurrent added to a reference current supplied by the first currentcircuit to the third current circuit.

A filter circuit of the present invention is a first-order filtercircuit configured using the transconductance amplifiers describedabove, wherein:

the filter circuit comprises the transconductance amplifier and acapacitance, and is made up of a first and a second transconductanceamplifier, the first transconductance amplifier having an outputterminal and an inverting output terminal connected to an input terminaland an inverting input terminal of the second transconductanceamplifier, respectively, and grounded through capacitances, the secondtransconductance amplifier having an output terminal and an invertingoutput terminal connected to the inverting input terminal and an inputterminal of the second transconductance amplifier.

A filter circuit according to another aspect of the present invention isa fourth-order filter circuit configured using the transconductanceamplifiers described above, wherein:

the filter circuit comprises a first to a fourth transconductanceamplifier, the first transconductance amplifier having an outputterminal and an inverting output terminal connected to an input terminaland an inverting input terminal of the second transconductance amplifierand grounded through capacitances, the second transconductance amplifierhaving an output terminal and an inverting output terminal connected tothe input terminal and an inverting input terminal of the secondtransconductance amplifier and grounded through capacitances, the thirdtransconductance amplifier having an output terminal and an invertingoutput terminal connected to an inverting input terminal and an inputterminal of the third transconductance amplifier, the fourthtransconductance amplifier having an input terminal and an invertinginput terminal connected to the output terminal and inverting outputterminal of the second transconductance amplifier, the fourthtransconductance amplifier having an output terminal and an invertingoutput terminal connected to the inverting input terminal and inputterminal of the second transconductance amplifier.

A filter circuit according to a further aspect of the present inventioncomprises one first-order filter circuit described above connected inseries with two fourth-order filter circuits described above.

A voltage generating circuit according to the present invention is avoltage generating circuit configured using the transconductanceamplifier described above, characterized in that:

the transconductance amplifier has an output terminal and an invertingoutput terminal connected to an inverting input terminal and an inputterminal, and

the voltage generating circuit comprises a capacitance for alternatinglygrounding one output part of the transconductance amplifier.

A voltage generating circuit according to another aspect of the presentinvention is a voltage generating circuit configured using thetransconductance amplifier described above, characterized in that:

the voltage generating circuit comprises a first and a secondtransconductance amplifier and a capacitance, the first transconductanceamplifier having an output terminal and an inverting output terminalconnected to an input terminal and an inverting input terminal of thesecond transconductance amplifier and connected to an inverting inputterminal and an input terminal of the first transconductance amplifier,the input terminal and inverting input terminal of the firsttransconductance amplifier connected to an input through capacitances,respectively, an output terminal and an inverting output terminal of thesecond transconductance amplifier respectively serving as outputs.

A current controlled oscillator of the present invention is a currentcontrolled oscillator configured using the voltage generating circuitdescribed above, which comprises:

a plurality of resistors provided in series between a power supply and aground;

a switch group provided between the plurality of resistors and an inputof the voltage generating circuit for selectively applying a voltagedivided by the plurality of resistors to the voltage generating circuit;

a first and a second comparator for comparing terminal voltages of theplurality of resistors provided in series with an output of the voltagegenerating circuit; and

a flip-flop whose state changes in accordance with outputs of the firstand second comparators and which generates an output which defines anoscillation frequency and is used as a switching control signal for theswitch group.

A PLL circuit of the present invention is a PLL circuit configured usingthe current controlled oscillator described above, which comprises:

a current controlled oscillator, whose oscillation frequency iscontrolled by a current control signal;

a phase detector for receiving a reference frequency signal and anoutput of the current controlled oscillator to generate a signal inaccordance with a phase difference therebetween; and

a voltage/current converter for converting an output of the phasedetector to a current, and supplying the same to the control signalinput terminal of the current controlled oscillator.

In the present invention configured as described above, the common-modecurrent generating circuit generates a current of an common-modecomponent alone. This current of the common-mode component isdistributed by the first current mirror circuit, and is subtracted fromthe output of each voltage/current converting element, thereby leavingonly the current of a differential component at each output. In thisevent, while the error component of an amount depending on thecommon-mode component generated by the first current mirror circuit isadded to each output, these error components are removed by the secondcurrent mirror circuit and third current mirror circuit.

A transconductance amplifier according to another aspect of the presentinvention is a transconductance amplifier for generating a first outputvoltage signal and a second output voltage signal proportional to thedifference between a first input voltage signal and a second inputvoltage signal applied thereto from a first and a second output stage,respectively. The transconductance amplifier is characterized bycomprising:

a feedback signal output terminal for outputting common-mode componentsof the first output voltage signal and second output voltage signal;

a feedback signal input terminal; and

a reference signal input terminal,

provided in the first and second output stages, respectively; and

feedback signal conveying means for controlling the first output voltagesignal or second output voltage signal in accordance with an inputsignal to the feedback signal input terminal and an input signal to thereference signal input terminal,

wherein the feedback signal conveying means is connected to each outputstage.

In this event, the feedback signal conveying means may comprise:

a current mirror circuit having an output part connected to the outputstage;

a first transistor of a first conductivity type having a controlterminal connected to the feedback signal input terminal; and

a second transistor of a second conductivity type having a controlterminal connected to the reference signal input terminal fordetermining a reference current for the current mirror circuit togetherwith the first transistor.

A transconductance amplifier according to a second aspect of the presentinvention is a transconductance amplifier for generating a first outputvoltage signal and a second output voltage signal proportional to thedifference between a first input voltage signal and a second inputvoltage signal applied thereto from a first and a second output stage,respectively. The transconductance amplifier is characterized bycomprising:

a feedback signal output terminal for outputting common-mode componentsof the first output voltage signal and second output voltage signal;

a reference signal input terminal for receiving a reference signal forbringing each of the first and second output stages into a predeterminedbias state;

a feedback signal input terminal; and

feedback signal communicating means for controlling the first outputvoltage signal or the second output voltage signal in accordance with aninput signal to the feedback signal input terminal and an input signalto the reference signal input terminal,

provided in the first and second output stages, respectively; and

feedback signal conveying means for controlling the first output voltagesignal or second output voltage signal in accordance with an inputsignal to the feedback signal input terminal and an input signal to thereference signal input terminal,

wherein the feedback signal conveying means is connected to each outputstage.

In this event, the feedback signal conveying means may comprise:

a current mirror circuit having an output part connected to the outputstage;

a first transistor of a first conductivity type having a controlterminal connected to the feedback signal input terminal; and

a second transistor of a second conductivity type for receiving thefirst input voltage signal or the first input voltage signal at acontrol terminal to determine a reference current for the current mirrorcircuit together with the first transistor.

Also, the feedback signal conveying means may be a transistor which hasa control terminal connected to the feedback signal input terminal andan output part connected to the output stage.

A filter circuit of the present invention is a first-order filtercircuit configured using the transconductance amplifier described above,wherein:

the filter circuit comprises the transconductance amplifier and acapacitance, and is made up of a first and a second transconductanceamplifier, the first transconductance amplifier having an outputterminal and an inverting output terminal connected to an input terminaland an inverting input terminal of the second transconductanceamplifier, respectively, and grounded through capacitances, the secondtransconductance amplifier having an output terminal and an invertingoutput terminal connected to the inverting input terminal and inputterminal of the second transconductance amplifier.

A filter circuit according to another aspect of the present invention isa fourth-order filter circuit configured using the transconductanceamplifiers described above, wherein:

the filter circuit comprises a first to a fourth transconductanceamplifier, the first transconductance amplifier having an outputterminal and an inverting output terminal connected to an input terminaland an inverting input terminal of the second transconductance amplifierand grounded through capacitances, the second transconductance amplifierhaving an output terminal and an inverting output terminal connected tothe input terminal and inverting input terminal of the secondtransconductance amplifier and grounded through capacitances, the thirdtransconductance amplifier having an output terminal and an invertingoutput terminal connected to an inverting input terminal and an inputterminal of the third transconductance amplifier, the fourthtransconductance amplifier having an input terminal and an invertinginput terminal connected to the output terminal and inverting outputterminal of the second transconductance amplifier, the fourthtransconductance amplifier having an output terminal and an invertingoutput terminal connected to the inverting input terminal and inputterminal of the second transconductance amplifier.

A filter circuit according to a further aspect of the present inventioncomprises one first-order filter circuit described above connected inseries with two fourth-order filter circuits described above.

A voltage generating circuit of the present invention is a voltagegenerating circuit configured using the transconductance amplifierdescribed in any of the foregoing, characterized in that:

a feedback signal input terminal is used as a control signal inputterminal for generating a bias current to change mutual conductance, and

the voltage generating circuit comprises a capacitance for alternatinglygrounding an output current.

A current controlled oscillator of the present invention is a currentcontrolled oscillator configured using the voltage generating circuitdescribed above, comprising:

a plurality of resistors provided in series between a power supply and aground;

a switch group provided between the plurality of resistors and an inputof the voltage generating circuit for selectively applying a voltagedivided by the plurality of resistors to the voltage generating circuit;

a first and a second comparator for comparing terminal voltages of theplurality of resistors provided in series with an output of the voltagegenerating circuit; and

a flip-flop whose state changes in accordance with outputs of the firstand second comparators and which generates an output which defines anoscillation frequency and is used as a switching control signal for theswitch group.

A PLL circuit of the present invention is a PLL circuit configured usingthe current controlled oscillator described above, comprising:

a current controlled oscillator, whose oscillation frequency iscontrolled by a current control signal;

a phase detector for receiving a reference frequency signal and anoutput of the current controlled oscillator to generate a signal inaccordance with a phase difference therebetween; and

a voltage/current converter for converting an output of the phasedetector to a current, and supplying the same to the control signalinput terminal of the current controlled oscillator.

EFFECTS OF THE INVENTION

In the present invention, since the gain of the common-mode component isreduced in the output, it is possible to effectively provide asufficient amplitude and to increase the degree of freedom of design.

Also, in the present invention configured as described above, the outputstage is connected only to the feedback signal conveying means, morespecifically, to the output part of the current mirror circuit ortransistor, so that only one transistor is connected other than thetransistor which forms part of the output stage of the transconductanceamplifier. Accordingly, a smaller number of transistors is connectedthan before, making it possible to accomplish an OTA which comprises aCMFB circuit that restrains degradations in characteristics.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1

A diagram showing the basic configuration of a transconductanceamplifier.

FIG. 2

A circuit diagram showing a specific configuration of a degenerateddifferential OTA which is an OTA, whose mutual conductance G_(m) ofwhich is controlled.

FIG. 3 a

A circuit diagram showing the configuration of a pseudo differentialinput type OTA.

FIG. 3 b

An equivalent circuit diagram showing the configuration of an outputstage of the pseudo differential input type OTA.

FIG. 4

A diagram showing the pseudo differential input type OTA shown in FIG. 3in a functional block form.

FIG. 5 a

A block diagram conceptually showing the configuration of a CMFBcircuit.

FIG. 5 b

A circuit diagram showing a specific configuration of a CMFB circuit.

FIG. 5 c

A block diagram showing an exemplary application of a CMFB circuit.

FIG. 6

An equivalent circuit diagram showing the configuration of a firstembodiment according to the present invention.

FIG. 7

A circuit diagram showing the configuration of the first embodimentaccording to the present invention.

FIG. 8

A diagram for describing effects of the first embodiment according tothe present invention.

FIG. 9

A diagram for describing effects of the first embodiment according tothe present invention.

FIG. 10

A diagram for describing effects of the first embodiment according tothe present invention.

FIG. 11

A diagram for describing effects of the first embodiment according tothe present invention.

FIG. 12

A diagram for describing effects of the first embodiment according tothe present invention.

FIG. 13

A diagram for describing effects of the first embodiment according tothe present invention.

FIG. 14

A diagram for describing effects of the first embodiment according tothe present invention.

FIG. 15

A diagram for describing effects of the first embodiment according tothe present invention.

FIG. 16

A diagram for describing effects of the first embodiment according tothe present invention.

FIG. 17

A circuit diagram showing a first exemplary modification to the firstembodiment according to the present invention.

FIG. 18

A circuit diagram showing a second exemplary modification to the firstembodiment according to the present invention.

FIG. 19

An equivalent circuit diagram showing the configuration of a secondembodiment according to the present invention.

FIG. 20

A circuit diagram showing the configuration of the second embodimentaccording to the present invention.

FIG. 21

An equivalent circuit showing the configuration of a third embodimentaccording to the present invention.

FIG. 22

An equivalent circuit diagram showing the configuration of a fourthembodiment according to the present invention.

FIG. 23

A circuit diagram showing the configuration of the fourth embodimentaccording to the present invention.

FIG. 24

A circuit diagram showing the configuration of a fifth embodimentaccording to the present invention.

FIG. 25

A circuit diagram showing the configuration of a sixth embodimentaccording to the present invention.

FIG. 26

A circuit diagram showing the configuration of a seventh embodimentaccording to the present invention.

FIG. 27

A circuit diagram showing the configuration of an eighth embodimentaccording to the present invention.

FIG. 28

A circuit diagram showing the configuration of a ninth embodimentaccording to the present invention.

FIG. 29 a

A diagram showing a tenth embodiment according to the present invention.

FIG. 29 b

A circuit diagram showing the configuration of first-order filter 241 inFIG. 29 a.

FIG. 29 c

A circuit diagram showing the configuration of fourth-order filters 242,243 in FIG. 29 a.

FIG. 30 a

A block diagram showing the circuit configuration of a PLL of aneleventh embodiment according to the present invention.

FIG. 30 b

A circuit diagram showing the configuration of current controlledoscillator 255 in FIG. 30 a.

FIG. 30 c

A circuit diagram specifically showing the configuration of a comparisonvoltage generating circuit 257 shown in FIG. 30 b.

FIG. 31

A circuit diagram showing the configuration of a twelfth embodiment ofthe present invention.

FIG. 32

A circuit diagram showing the configuration of a thirteenth embodimentof the present invention.

FIG. 33

A circuit diagram showing the configuration of a fourteenth embodimentof the present invention.

FIG. 34 a

A block diagram showing the configuration of a filter of a fifteenthembodiment according to the present invention.

FIG. 34 b

A circuit diagram showing the configuration of first-order filter 241 inFIG. 34 a.

FIG. 34 c

A circuit diagram showing the configuration of fourth-order filters 242,243 in FIG. 34 a.

FIG. 35 a

A block diagram showing the circuit configuration of a pLL of asixteenth embodiment according to the present invention.

FIG. 35 b

A circuit diagram showing the configuration of current controlledoscillator 255 in FIG. 35 a.

DESCRIPTION OF REFERENCE NUMERALS

-   101 First Voltage/Current Converting Element-   102 Second Voltage/Current Converting Element-   103 Third Voltage/Current Converting Element-   104 Fourth Voltage/Current Converting Element-   105 Fifth Voltage/Current Converting Element-   106 Sixth Voltage/Current Converting Element-   107 In-Phase Current Generating Part-   108 First Current Mirror Circuit-   109 Second Current Mirror Circuit-   110 Third Current Mirror Circuit-   M_(03A), M_(3A), M_(5A), M_(6A), M_(7A), M_(03B), M_(3B), M_(5B),    M_(6B), M_(7B) n-mos Transistors-   M_(04A), M_(4A), M_(8A), M₀₄, M_(4B), M_(8B) p-mos Transistors

BEST MODE FOR CARRYING OUT THE INVENTION

Next, embodiments of the present invention will be described withreference to the drawings.

First Embodiment

FIGS. 6 and 7 are diagrams showing the configuration of a firstembodiment according to the present invention, where FIG. 6 is anequivalent circuit diagram, and FIG. 7 is a circuit diagram.

As shown in the equivalent circuit of FIG. 6, this embodiment comprisesfirst to sixth voltage/current converting elements 101-106, common-modecurrent generating part 107, first to third current mirror circuits108-110.

Replacing the configuration of FIG. 6 with a specific circuit shown inFIG. 7, first to sixth voltage/current converting elements 101-106 aremade up of n-mos transistors M_(1F), M_(1C), M_(1B), M_(1E), M_(1D),M_(1A). The first current mirror circuit is made up of p-mos transistorsM_(2A)-M_(2F). The second current mirror circuit is made up of n-mostransistors M_(3C), M_(3D), while the third current mirror circuit ismade up of n-mos transistors M_(3A), M_(3B). Also, common-mode currentgenerating part 107 is made up of second voltage/current convertingelement 102 (M_(1C)) and fifth voltage/current converting element 105(M_(1D)).

With regard to the size of the transistors, in the first embodiment,transistors M_(1A)-M_(1F) are identical in size to one another, buttransistors M_(2A)-M_(2F) may differ in size from transistorsM_(1A)-M_(1F). Also, transistors M_(3A), M_(3B) may differ in size. Inthis regard, the size of each of transistors M_(1A)-M_(1F) may bechanged to a different size depending on a particular usage as in theexemplary modifications shown below.

Sources and gates of p-mos transistors M_(2A)-M_(2F) are made common,and the sources are connected to a power supply. Each drain of p-mostransistors M_(2A)-M_(2F) is connected to each drain of n-mostransistors M_(1F), M_(1C), M_(1A), M_(1B), M_(1E), M_(1A). Sources ofn-mos transistors M_(1A)-M_(1F), M_(3A)-M_(3D) are grounded. N-mostransistors M_(3C), M_(3D) which form part of second current mirrorcircuit have their drains connected to the drains of p-mos transistorsM_(2E), M_(2F), and have their gates connected commonly to the drain ofn-mos transistor M_(3C). N-mos transistors M_(3A), M_(3B) which formpart of the third current mirror circuit have their drains connected tothe drains of p-mos transistors M_(2A), M_(2B), and have their gatesconnected in common to the drain of n-mos transistor M_(3B). N-mostransistors M_(1C), M_(1D) which form part of common-mode currentgenerating part 107 have their respective gates connected to respectivegates of n-mos transistors M_(1B), M_(1E), respectively, and have theirdrains connected commonly to each gate of p-mos transistorsM_(2A)-M_(2F).

In this regard, transistors have their drain, gate, and sourceconnected, respectively, for example, p-mos transistors M_(2C) andM_(2D) can be separately provided as shown in FIG. 7, these transistorsmay be integrated into one by increasing the size twice. When p-mostransistors M_(2C) and M_(2D) are integrated into one, the first currentmirror circuit is made up of five p-mos transistors.

In the circuit configured as described above, signal voltages V_(a) andV_(b) are converted to currents by the first to sixth voltage/currentconverting elements. Common-mode current generating circuit 107generates a current proportional to an common-mode component of an inputsignal. The first current mirror circuit inverts the characteristic ofthis current proportional to the common-mode component, and subtractsthe same from outputs of the first, third, fourth, and sixthvoltage/current converting elements. Among the subtracted signals, thepolarity of one is inverted by the second and third current mirrorcircuits such that the one signal is subtracted from the other togenerate outputs I_(OUT1), I_(OUT2).

In this embodiment, signals converted from voltage to currentcorresponding to signal voltages V_(a) and V_(b) include a current of ancommon-mode component and a current of a differential component addedthereto. In the common-mode current generating circuit, the outputs ofthe second and fifth voltage/current converting elements areshort-circuited, so that the current of the differential component isremoved, with the current of the common-mode component alone beingoutput. This current of the common-mode component is distributed by thefirst current mirror circuit, and subtracted from outputs of the first,fourth, third, and sixth voltage/current converting elements, therebycausing each output to comprise the current of the differentialcomponent alone. However, generally, a circuit made up of transistorshas transistors with limited output conductance, and thus the errorcomponent of an amount depending on the common-mode component is addedto the output of the first current mirror circuit in this embodiment inaddition to the current of the common-mode component. Consequently, theerror component of the same amount is added to all of the outputs of thefirst, fourth, third, and sixth voltage/current converting elements. Inthis embodiment, the error components included in the outputs of thefirst and fourth voltage/current converting elements are removed by thesecond current mirror circuit, while the error components included inthe outputs of the third and sixth voltage/current converting elementsare removed by the third current mirror circuit. These outputs free fromthe error components are labeled output I_(OUT1), output I_(OUT2),respectively. Accordingly, in this embodiment, the error componentsattributable to the common-mode components can be reduced to providesufficient amplitude.

The foregoing effect will be described by calculating the gain of thecommon-mode component.

First, a diode model is analyzed. FIG. 8 shows equivalent circuits of an-mos transistor and a P-mos transistor, where the relationship of thecurrent is represented by the following Equation (1):

$\begin{matrix}\left\lbrack {{Equation}\mspace{14mu} 1} \right\rbrack & \; \\{{{{{- g_{o}}v_{x}} - {g_{m}v_{x}} + I_{x}} = 0}{Z_{M} = {\frac{V_{x}}{I_{x}} = \frac{1}{g_{m} + g_{o}}}}} & (1)\end{matrix}$

where g_(m), g_(o) are mutual conductance and output conductance of thetransistor, respectively. In the following, the mutual conductance andoutput conductance of a transistor are labeled g_(m), g_(o) with asuffix added thereto, unless otherwise noted.

FIG. 9 is an equivalent circuit of a source-grounded circuit with adiode added thereto, where the relationship between input voltage v_(i)and output voltage v_(o) is represented by the following Equation (2):

$\begin{matrix}\left\lbrack {{Equation}\mspace{14mu} 2} \right\rbrack & \; \\{{{{g_{m\; 1}v_{i}} + {g_{o\; 1}v_{o}} + {\left( {g_{m\; 2} + g_{o\; 2}} \right)v_{o}}} = 0}{\frac{v_{o}}{v_{i}} = {- \frac{g_{m\; 1}}{\left( {g_{o\; 1} + g_{o\; 2} + g_{m\; 2}} \right)}}}} & (2)\end{matrix}$

FIG. 10 is a circuit diagram showing only a one-side portion of thecircuit shown in FIG. 3, and FIG. 11 is an equivalent circuit diagramthereof.

In the circuit shown in FIG. 10, the relationship between input(common-mode input) V_(i) and output V₀₁ is represented by Equation (3)shown below:

$\begin{matrix}\left\lbrack {{Equation}\mspace{14mu} 3} \right\rbrack & \; \\{v_{o\; 1} = {{- \frac{g_{m\; 1}}{g_{o\; 1} + g_{o\; 2} + g_{m\; 2}}}v_{i}}} & (3)\end{matrix}$

where g_(m1) represents mutual conductance of M_(1A), M_(01A), andg_(m2) represents mutual conductance of M_(2A), M_(02A), respectively,while g₀₁ represents output conductance of M_(1A), M_(01A), and g₀₂represents output conductance of M_(2A), M_(02A), respectively.

From the equivalent circuit shown in FIG. 11 and Equation (3), theequation is expanded to derive the gain:

$\begin{matrix}\left\lbrack {{Equation}\mspace{14mu} 4} \right\rbrack & \; \\{{{{g_{m\; 1}v_{i}} + {g_{m\; 2}v_{o\; 1}} + {g_{o\; 1}v_{o\; 2}} + {g_{o\; 2}v_{o\; 2}}} = 0}{{{g_{m\; 1}v_{i}} - {\frac{g_{m\; 1}g_{m\; 2}}{g_{o\; 1} + g_{o\; 2} + g_{m\; 2}}v_{i}} + {\left( {g_{o\; 1} + g_{o\; 2}} \right)v_{o\; 2}}} = 0}{{{v_{i}\left( {g_{m\; 1} - \frac{g_{m\; 1}g_{m\; 2}}{g_{o\; 1} + g_{o\; 2} + g_{m\; 2}}} \right)} + {\left( {g_{o\; 1} + g_{o\; 2}} \right)v_{o\; 2}}} = 0}{{{v_{i}\left( \frac{{g_{o\; 1}g_{m\; 1}} + {g_{o\; 2}g_{m\; 1}} + {g_{m\; 1}g_{m\; 2}} - {g_{m\; 1}g_{m\; 2}}}{g_{o\; 1} + g_{o\; 2} + g_{m\; 2}} \right)} + {\left( {g_{o\; 1} + g_{o\; 2}} \right)v_{o\; 2}}} = 0}{{\left( \frac{g_{o\; 1} + g_{o\; 2}}{g_{o\; 1} + g_{o\; 2} + g_{m\; 2}} \right)v_{i}g_{m\; 1}} = {{- \left( {g_{o\; 1} + g_{o\; 2}} \right)}v_{o\; 2}}}{\frac{v_{o\; 2}}{v_{i}} = {- \frac{g_{m\; 1}}{\left( {g_{o\; 1} + g_{o\; 2} + g_{m\; 2}} \right)}}}} & (4)\end{matrix}$

Here, mutual conductance g_(m) of a transistor is generally larger thanoutput conductance g0 by a factor of 10 to 100, so that when Equation(4) is simplified on the assumption that g_(m)>>g_(o) is satisfied, theresulting equation is represented as follows:

$\begin{matrix}\left\lbrack {{Equation}\mspace{14mu} 5} \right\rbrack & \; \\{\frac{v_{o\; 2}}{v_{i}} \approx {- \frac{g_{m\; 1}}{g_{m\; 2}}}} & (5)\end{matrix}$

From the foregoing, in the example shown in FIG. 3, when an equal valueis selected for mutual conductance g_(m1), g_(m2) of transistors M_(1A),M_(01A), M_(2A), M_(02A), the common-mode gain of the output withrespect to the input signal is approximately −1 time from Equation (5).Also, supposing that values are selected for mutual conductance g_(m1),m2 of transistors M_(1A), M_(01A), M_(2A), M_(02A) such that Equation(2) derives − 1/10, mutual conductance g_(m) is proportional to the sizeof the transistors, so that the shapes of p-oh transistors and n-ohtransistors must be at 1:10. While the size of a transistor isdetermined in consideration of a reduction in source voltage, a noisemargin, variations in performance of the transistor, and the like, thedesign becomes further difficult due to the ratio as mentioned abovewhich is included in the condition.

Next, the OTA of this embodiment is analyzed. FIG. 12 is a circuitdiagram showing only one-side portion of the circuit shown in FIG. 7.For purposes of analysis, transistors M_(2C), M_(3B) are replaced withloads Z_(M2), Z_(M3), as shown in FIG. 13.

First, v_(o1) is derived. By considering the equivalent circuit shown inFIG. 14 in a manner similar to the derivation of Equation (2), theinput/output relationship is represented as follows:

$\begin{matrix}\left\lbrack {{Equation}\mspace{14mu} 6} \right\rbrack & \; \\{v_{o} = {{- \frac{g_{m\; 1}}{\left( {g_{o\; 1} + g_{o\; 2} + g_{m\; 2}} \right)}}v_{i}}} & (6)\end{matrix}$

Next, v_(o2) is derived. Taking into account the equivalent circuitshown in FIG. 15 in which Z_(M3), the input/output relationship isrepresented as follows:

$\begin{matrix}\left\lbrack {{Equation}\mspace{14mu} 7} \right\rbrack & \; \\{{0 = {{g_{m\; 1}v_{i}} - {\frac{g_{m\; 1}g_{m\; 2}}{\left( {g_{o\; 1} + g_{o\; 2} + g_{m\; 2}} \right)}v_{i}} + {\left( {g_{m\; 3} + g_{o\; 3}} \right)v_{o\; 2}} + {g_{o\; 1}v_{o\; 2}} + {g_{o\; 2}v_{o\; 2}}}}{{v_{i}\left( {\frac{g_{m\; 1}g_{m\; 2}}{g_{o\; 1} + g_{o\; 2} + g_{m\; 2}} - g_{m\; 1}} \right)} = {\left( {g_{o\; 1} + g_{o\; 2} + g_{o\; 3} + g_{m\; 3}} \right)v_{o\; 2}}}{v_{o\; 2} = {\left( \frac{\frac{g_{m\; 1}g_{m\; 2}}{g_{o\; 1} + g_{o\; 2} + g_{m\; 2}} - g_{m\; 1}}{g_{o\; 1} + g_{o\; 2} + g_{o\; 3} + g_{m\; 2}} \right)v_{i}}}} & (7)\end{matrix}$

Next, vo3 is derived using vo1, vo2 which was calculated in theforegoing manner. The equivalent circuit is, as shown in FIG. 16, andthe input/output relationship is represented as follows:

[Equation  8]0 = g_(m 1)v_(i) + g_(m 3)v_(o 2) + g_(m 2)v_(o 1) + g_(o 1)v_(o 3) + g_(o 2)v_(o 3) + g_(o 3)v_(o 3)$0 = {{g_{m\; 1}v_{i}} + {\left( \frac{\frac{g_{m\; 1}g_{m\; 2}}{g_{o\; 1} + g_{o\; 2} + g_{m\; 2}} - g_{m\; 1}}{g_{o\; 1} + g_{o\; 2} + g_{o\; 3} + g_{m\; 3}} \right)g_{m\; 3}v_{i}} - {\left( \frac{g_{m\; 1}}{g_{o\; 1} + g_{o\; 2} + g_{m\; 2}} \right)g_{m\; 2}v_{i}} + {\left( {g_{o\; 1} + g_{o\; 2} + g_{o\; 3}} \right)v_{o\; 3}}}$$0 = {{\left( {1 + \frac{\frac{g_{m\; 2}g_{m\; 3}}{g_{o\; 1} + g_{o\; 2} + g_{m\; 2}} - g_{m\; 3}}{g_{o\; 1} + g_{o\; 2} + g_{o\; 3} + g_{m\; 3}} - \frac{g_{m\; 2}}{g_{o\; 1} + g_{o\; 2} + g_{m\; 2}}} \right)g_{m\; 2}v_{i}} + {\left( {g_{o\; 1} + g_{o\; 2} + g_{o\; 3}} \right)v_{o\; 3}}}$

When g_(o1)+g_(o2)=A,

[Equation  9]$0 = {{\left( {1 + \frac{\frac{g_{m\; 2}g_{m\; 3}}{A + g_{m\; 2}} - g_{m\; 3}}{A + g_{o\; 3} + g_{m\; 3}} - \frac{g_{m\; 2}}{A + g_{m\; 2}}} \right)g_{m\; 1}v_{i}} + {\left( {A + g_{o\; 3}} \right)v_{o\; 3}}}$$0 = {{\left( {1 + \frac{- {Ag}_{m\; 3}}{\left( {A + g_{m\; 2}} \right)\left( {A + g_{o\; 3} + g_{m\; 3}} \right)} - \frac{g_{m\; 2}}{A + g_{m\; 2}}} \right)g_{m\; 1}v_{i}} + {\left( {A + g_{o\; 3}} \right)v_{o\; 3}}}$$0 = \left( \frac{\begin{matrix}{A^{2} + {Ag}_{o\; 3} + {Ag}_{m\; 3} + {Ag}_{m\; 2} + {g_{m\; 2}g_{o\; 3}} + {g_{m\; 2}g_{m\; 3}} -} \\{{Ag}_{m\; 3} - {Ag}_{m\; 2} - {g_{m\; 2}g_{o\; 3}} - {g_{m\; 2}g_{m\; 3}}}\end{matrix}}{\left( {A + g_{m\; 2}} \right)\left( {A + g_{o\; 3} + g_{m\; 2}} \right)} \right)$g_(m 1)v_(i) + (A + g_(o 3))v_(o 3)$0 = {{\left( \frac{A\left( {A + g_{o\; 3}} \right)}{\left( {A + g_{m\; 2}} \right)\left( {A + g_{o\; 3} + g_{m\; 3}} \right)} \right)g_{m\; 1}v_{i}} + {\left( {A + g_{o\; 3}} \right)v_{o\; 3}}}$$v_{o\; 3} = {{- \frac{{Ag}_{m\; 1}}{\left( {A + g_{m\; 2}} \right)\left( {A + g_{o\; 3} + g_{m\; 3}} \right)}}v_{i}}$

A substitution of A results in the following equation:

$\begin{matrix}\left\lbrack {{Equation}\mspace{14mu} 10} \right\rbrack & \; \\{v_{o\; 3} = {{- \frac{\left( {g_{o\; 1} + g_{o\; 2}} \right)g_{m\; 1}}{\left( {g_{o\; 1} + g_{o\; 2} + g_{m\; 2}} \right)\left( {g_{o\; 1} + g_{o\; 2} + g_{o\; 3} + g_{m\; 3}} \right)}}v_{i}}} & (8)\end{matrix}$

When g_(o1)+g_(o2)<<g_(m2) and g_(o1)+g_(o2)+g_(o3)<<g_(m3),

$\begin{matrix}\left\lbrack {{Equation}\mspace{14mu} 11} \right\rbrack & \; \\{v_{o\; 3} \cong {{- \frac{g_{m\; 1}}{g_{m\; 2}g_{m\; 3}}}\left( {g_{o\; 1} + g_{o\; 2}} \right)v_{i}}} & (9)\end{matrix}$

$\begin{matrix}\left\lbrack {{Equation}\mspace{14mu} 12} \right\rbrack & \; \\{\frac{v_{o\; 3}}{v_{i}} \cong {{- \frac{g_{m\; 1}}{g_{m\; 2}g_{m\; 3}}}\left( {g_{o\; 1} + g_{o\; 2}} \right)}} & (10)\end{matrix}$

where g_(m1), g_(m2), g_(m3) represent the mutual conductance oftransistors M_(1A), M_(1B), M_(1C), M_(2A), M_(2B), M_(2C), M_(3A),M_(3B), and g01, g02, g03 represent the output conductance of M_(1A),M_(1B), M_(1C), M_(2A), M_(2B), M_(2C), M_(3A), M_(3B). Generally, themutual conductance of transistors is larger than the output conductanceby a factor of 10 to 100, Equation (9) can be simplified as Equation(10).

Here, for comparing the circuit of this embodiment with the related artdescribed with reference to FIG. 3, g_(m1), g_(m2) of the transistors inFIG. 3 and the transistors in FIG. 7 are made the same, and the ratio ofEquation (4) to Equation (8) is taken as follows:

[Equation  13]$\frac{\frac{v_{o\; 3}}{v_{i}}}{\frac{v_{o\; 2}}{v_{i}}} = {\frac{\left( {g_{o\; 1} + g_{o\; 2}} \right)g_{m\; 1}}{\left( {g_{o\; 1} + g_{o\; 2} + g_{m\; 2}} \right)\left( {g_{o\; 1} + g_{o\; 2} + g_{o\; 3} + g_{m\; 3}} \right)}/\frac{g_{m\; 1}}{\left( {g_{o\; 1} + g_{o\; 2} + g_{m\; 2}} \right)}}$$\frac{\frac{v_{o\; 3}}{v_{i}}}{\frac{v_{o\; 2}}{v_{i}}} = \frac{g_{o\; 1} + g_{o\; 2}}{g_{o\; 1} + g_{o\; 2} + g_{o\; 3} + g_{m\; 3}}$

$\begin{matrix}\left\lbrack {{Equation}\mspace{14mu} 14} \right\rbrack & \; \\{\frac{\frac{v_{o\; 3}}{v_{i}}}{\frac{v_{o\; 2}}{v_{i}}} \cong \frac{g_{o\; 1} + g_{o\; 2}}{g_{m\; 3}}} & (11)\end{matrix}$

As is apparent from Equation (11), assuming that the mutual conductanceg_(m) of the transistors is larger than output conductance g_(o) by afactor of 10 to 100, it is understood that the gain of the common-modecomponent can be reduced to ⅕ to 1/50. Accordingly, when the circuitconfiguration of this embodiment is used, the design can be made withoutpaying any concern to limiting the size of transistors due to thecommon-mode gain, resulting in the design with a higher degree offreedom.

While this embodiment has been described on the assumption that thetransistors which form part of the circuit comprise p-mos transistorsand n-mos transistors, the circuit may be made up of JFETs and bipolartransistors. Also, the p-mos transistors shown in FIG. 7 may be replacedwith n-mos transistors, and the n-mos transistors may be replaced withp-mos transistors. This applies to other embodiments.

(Exemplary Modification 1 to First Embodiment)

FIG. 17 is a circuit diagram showing an exemplary modification to thefirst embodiment. FIG. 17 differs from the first embodiment in that thesecond voltage/current converting element and fifth voltage/currentconverting element, which form part of the common-mode currentgenerating part, are smaller than the remaining voltage/currentconverting elements. Here, the size of transistors M_(1C) and M_(1D)(=W/L) is chosen to be a, while the size of transistors M_(1A), M_(1B),M_(1E), M_(1F) is chosen to be 2a. While the size of transistors M_(3A),M_(3B), M_(3C), M_(3D) can be determined irrespective of transistorsM_(1C), M_(1D) and transistors M_(1A), M_(1B), M_(1E), M_(1F), the sizeis herein chosen to be 2a.

Likewise, the size of transistors M_(2C), M_(2D) which forms part of thefirst current mirror circuit is also chosen to be a, and the size oftransistors M_(2A), M_(2B), M_(2E), M_(2F) is 2a. In other words, theratio of the size of transistors M_(1C) and M_(1D) which make up thecommon-mode current generating circuit to the size of transistorsM_(1A), M_(1B), M_(1E), M_(1F), which are the remaining voltage/currentconverting elements, is set to 1:2, and within the first mirror circuit,the ratio of the size of transistors M_(2C), M_(2D) connected totransistor M_(1C), M_(1D) to the size of remaining transistors M_(2A),M_(2B), M_(2E), M_(2F) is also set to 1:2 in a similar manner.

Such a configuration can reduce the current flowing through common-modecurrent generating part 107, making it possible to reduce the inputimpedance. Also, the power consumption can be reduced by reducingtransistors not related to transistors which make up the output stage(M_(1A), M_(2A), M_(3A), M_(1F), M_(2F), M_(3D)).

Also, in this embodiment, the ratio of the sizes (transistors M_(1C),M_(1D)):(transistors M_(1A), M_(1B), M_(1E), M_(1F), M_(3A), M_(3B),M_(3C), M_(3D)) is set to 1:2, but transistors of any size may be usedas long as 1:n (n>1). Also, the size of M_(3A), M_(3B), M_(3C), M_(3D)can be independently set.

(Exemplary Modification 2 to First Embodiment)

FIG. 18 is a circuit diagram showing an exemplary modification to thefirst embodiment. FIG. 18 differs from the first embodiment in that thesize of transistors M_(1C), M_(1B), M_(1D), M_(1E) is chosen to be a,while the size of transistors M_(1A), M_(1F) is chosen to be 2a, so thatthey are at a ratio of 2:1; the size of transistors M_(3B), M_(3C) ischosen to be b, while the size of transistors M_(3A), M_(3D) is chosento be 2b, so that they are at a ratio of 2:1; and the size oftransistors M_(2B), M_(2D), M_(2B), M_(02E) is chosen to be c, while thesize of transistors M_(2A), M_(2F) is chosen to be 2c, so that they arein a ratio of 1:2.

With such a configuration, part of transistors can be reduced in area,thus making it possible to produce an effect of reducing the overallcircuit area, while having similar effects to those of the firstembodiment. Also, in this embodiment, the ratio of the sizes is set to2:1, but transistors of any size may be used as long as n:1 (n>1), in amanner similar to Exemplary Modification 1 to the first embodiment.Also, the sizes of the transistors can be changed in a similar manner.

Alternatively, a well separation type, which is effective in reducingnoise, can be used for the n-mos transistors.

The foregoing contents related to the transistors are similar in eachembodiment described below.

Second Embodiment

FIGS. 19 and 20 are diagrams showing the configuration of a secondembodiment, where FIG. 19 is an equivalent circuit, and FIG. 20 is acircuit diagram.

This embodiment comprises seventh voltage/current converting element 401in common-mode current generating part 107 in the first embodiment toprovide common-mode current generating part 107′. In the circuit diagramspecifically shown in FIG. 20, seventh voltage/current convertingelement 401 comprises source-grounded n-mos transistor M₄. N-mostransistor M₄ has a drain connected in common to each gate of p-mostransistors M_(2A)-M_(2F), and a gate applied with bias voltage V_(c).

In the first embodiment, when an error component included in an outputcurrent of first current mirror circuit 108 is very small, the currentflowing through transistors M_(2A)-M_(2F) is substantially the same asthe current flowing through transistors M_(1A)-M_(1F). In this state, nobias current flows into transistors M_(3A)-M_(3D) which make up secondcurrent mirror circuit 109 and third current mirror circuit 110, butinstead the current flows into transistors M_(3A)-M_(3D) only in halfwaves of a signal, causing a loss in the signal.

In this embodiment, the seventh voltage/current converting element isprovided at an input of first current mirror circuit 108 to supply theinput of first current mirror circuit 108 with DC voltage V_(c) notrelated to a signal, thus making it possible to cause the currentflowing through transistors M_(2A)-M_(2F) to be larger than the currentflowing through transistors M_(1A)-M_(1F) at all times. DC currentV_(c), called herein, may be of any magnitude as long as it has a valuefrom a grounding point to a source voltage. As a result, it is possibleto solve a problem in which a bias current experiences difficulties inflowing into transistors M_(3A)-M_(3D), which make up second currentmirror circuit 109 and third current mirror circuit 110, and to processa differential signal over a full wave rather than half wave, making itless likely to produce a loss in the signal.

Third Embodiment

FIG. 21 is an equivalent circuit showing the configuration of a thirdembodiment according to the present invention.

In the configuration of this embodiment, in addition to the OTA made upof first to sixth voltage/current converting elements 101-106,common-mode current generating part 107, and first to third currentmirror circuits 108-110 of the first embodiment shown in FIG. 6, an OTAhaving the same configuration as this is provided. Seventh to twelfthvoltage/current converting elements 101′-106′, common-mode currentgenerating part 107′, and fifth to seventh current mirror circuits108′-110′, respectively, operate in a similar manner to first to sixthvoltage/current converting elements 101-106, common-mode currentgenerating part 107, and first to third current mirror circuits 108-110.

In this embodiment, fourth current mirror circuit 601 is providedbetween third current mirror circuit 110 of the OTA which generatesI_(OUT1) and the output, and eighth current mirror 601′ is providedbetween sixth current mirror circuit 109′ of the OTA which generatesI_(OUT2) and the output, to remove common-mode components by eachcurrent mirror circuit, thus further improving the common-mode gainreduction effect.

Fourth Embodiment

FIGS. 22 and 23 are diagrams showing the configuration of a fourthembodiment according to the present invention, where FIG. 22 is anequivalent circuit diagram, and FIG. 23 is a circuit diagram.

This embodiment comprises eighth voltage/current converting element 701and ninth voltage/current converting element 702 added to the firstembodiment. In the circuit diagram specifically shown in FIG. 23, eighthvoltage/current converting element 701 and ninth voltage/currentconverting element 702 are shown as p-mos transistors M_(4B), M_(4A).P-mos transistor M_(4B) has a source connected to a power supply, and adrain connected to gates of M_(3C), M_(3D) which make up the secondcurrent mirror circuit, while p-mos transistor M_(4A) has a sourceconnected to the power supply, and a drain connected to gates of M_(3A),M_(3B) which make up the third current mirror circuit. Gates of p-mostransistors M_(4B), M_(4A) are applied with bias voltage V_(c).

As described above, in the first embodiment, when an error componentincluded in an output current of first current mirror circuit 108 isvery small, no bias current flows into transistors M_(3A)-M_(3D) whichmake up second current mirror circuit 109 and third current mirrorcircuit 110, but instead a current flows into transistors M_(3A)-M_(3D)only in half waves of a signal, causing a loss in the signal.

In this embodiment, eighth voltage/current converting element 701 andninth voltage/current converting element 702 are provided at inputs ofsecond current mirror circuit 109 and third current mirror circuit 110to supply the inputs of second current mirror circuit 109 and thirdcurrent mirror circuit 110 with DC voltage V_(c) not related to asignal, thereby making it possible to address the problem in which nobias current flows into transistors M_(3A)-M_(3D) which make up secondcurrent mirror circuit 109 and third current mirror circuit 110, and toprocess a differential signal over a full wave rather than half wave,making it less likely to produce a loss in the signal.

Fifth Embodiment

FIG. 24 is a circuit diagram showing the configuration of a fifthembodiment according to the present invention.

In this embodiment, first current mirror circuit 108 made up of p-mostransistors M_(2A)-M_(2F) in the first embodiment shown in FIG. 7 isdesigned in a two-stage configuration with p-mos transistorsM_(2A)′-M_(2F)′, identical in configuration to p-mos transistorsM_(2A)-M_(2F), provided between first current mirror circuit 108 and apower supply, for use as first current mirror circuit 108′.

Some transistors developed in recent years have low threshold voltages,and this embodiment employs such transistors as transistors which makeup the first current mirror circuit. By arranging the transistors at twostages, the output impedance can be increased, though the proportion ofthe amplitude width occupying in a source voltage becomes lower, thusimproving the amplification accuracy.

Sixth Embodiment

FIG. 25 is a circuit diagram showing the configuration of a sixthembodiment.

This embodiment is also based on the premise that it uses transistorswhich have low threshold voltages, as is the case with the fifthembodiment.

In this embodiment, second current mirror circuit 109 made up of n-mostransistors M_(3C), M_(3D) and third current mirror circuit 110 made upof n-mos transistors M_(3A), M_(3B) in the first embodiment shown inFIG. 7 are designed in a two-stage configuration with n-mos transistorsM_(3C)′, M_(3D)′, identical in configuration to n-mos transistorsM_(3C), M_(3D), and n-mos transistors M_(3A)′, M_(3B)′, identical inconfiguration to n-mos transistors M_(3A), M_(3B), provided betweensecond and third current mirror circuit 108, 109 and a ground, for useas second current mirror circuit 109′, and third current mirror circuit110′. Likewise, in this embodiment, the amplification accuracy isimproved as is the case with the fifth embodiment.

Seventh Embodiment

FIG. 26 is a circuit diagram showing the configuration of a seventhembodiment according to the present invention.

This embodiment is a combination of the fifth embodiment shown in FIG.24 with the sixth embodiment shown in FIG. 25, wherein first mirrorcircuit 108 shown in FIG. 7 is replaced with current mirror circuit 108′shown in FIG. 24, and second mirror circuit 109 and third current mirrorcircuit 110 shown in FIG. 7 are replaced with second current mirrorcircuit 109′ and third current mirror circuit 110′ shown in FIG. 25. Inthis embodiment, the effects of the fifth embodiment can besynergistically combined with the effects of the sixth embodiment.

While the foregoing fifth to seventh embodiments have been described inconnection with examples in which each current mirror circuit isdesigned in a two-stage configuration, the current mirror circuit can bedesigned in a more-stage configuration in accordance with a reduction inthreshold voltage, and may of course be in such a configuration.

Eighth Embodiment

FIG. 27 is a circuit diagram showing the configuration of an eighthembodiment.

In this embodiment, seventh voltage/current converting element 401,second current mirror circuit 109, and third current mirror circuit 110,which are made up of n-mos transistors in the second embodiment shown inFIG. 10, are replaced with seventh voltage/current converting circuit401′, second current mirror circuit 109′, and third current mirrorcircuit 110′ which are made up of p-mos transistors.

P-mos transistor M₄″, which is seventh voltage/current convertingelement 401′, which is applied at a gate with DC voltage V_(c) forgenerating a bias current, has a source connected to a power supply, anda drain connected to the gates of p-mos transistors M_(2A)-M_(2F) whichmake up the first current mirror circuit.

Each of p-mos transistors M_(3C)″, M_(3D)″ which make up second currentmirror circuit 109′ has a source connected to the power supply, and agate connected to the drains of p-mos transistor M2E and n-mostransistor M1E. P-mos transistor M_(3C)″ has a drain connected to thegate of each p-mos transistor M_(3C)″, M_(3D)″, while p-mos transistorM_(3D)″ has a drain connected to the drains of p-mos transistor M2F andn-mos transistor M1F.

Each of p-mos transistors M3A″, M3B″ which make up third current mirrorcircuit 110′ has a source connected to the power supply, and a gateconnected to the drains of p-mos transistor M_(2B) and n-mos transistorM1B. P-mos transistor M3B″ has a drain connected to a gate of each p-mostransistor M3A″, M3B″, while p-mos transistor M3A″ has a drain connectedto the drains of p-mos transistor M_(2A) and n-mos transistor M1A.

This embodiment configured as described above is similar to the secondembodiment in that seventh voltage/current converting element 107′ isprovided at an input of first current mirror circuit 108 to supply theinput of first current mirror circuit 108 with DC voltage V_(c) notrelated to a signal, thereby making it possible to cause a currentflowing through transistors M_(2A)-M2F to be larger than a currentflowing through transistors M1A-M1F at all times. As a result, it ispossible to solve a problem in which a bias current experiencesdifficulties in flowing into transistors M_(3A)-M_(3D) which make upsecond current mirror circuit 109′ and third current mirror circuit110′, and to process a differential signal over a full wave rather thanhalf wave, making it less likely to produce a loss in the signal.

Ninth Embodiment

FIG. 28 is a circuit diagram showing the configuration of a ninthembodiment according to the present invention.

In this embodiment, common-mode current generating part 107 in the firstembodiment shown in FIG. 7 is replaced with common-mode currentgenerating part 107″ which comprises a current mirror circuit, thuscausing it to have a function as the seventh voltage/current convertingelement shown in FIG. 9.

Common-mode current generating part 107″ is provided with n-mostransistors M_(4A)′, M_(4B)′ in addition to the configuration ofcommon-mode current generating part 107 shown in FIG. 7. Each of n-mostransistors M_(4A)′, M_(4B)′ has a source grounded, and a drain and agate supplied with DC voltage V_(c) connected to the gates of p-mostransistors M_(2A)-M_(2F) in common with the drains of n-mos transistorsM_(1C), M_(1D)′.

In this embodiment configured in the foregoing manner, by applyingreference current I_(ref) into the gates of n-mos transistors M_(4A)′,M_(4B)′, a current flowing through transistors M_(2A)-M_(2F) can be madelarger than a current flowing through transistors M_(1A)-M_(1F) at alltimes, as is the case with the second embodiment. As a result, it ispossible to solve a problem in which a bias current experiencesdifficulties in flowing into transistors M_(3A)-M_(3D) which make upthird current mirror circuit 110, and to process a differential signalover a full wave rather than half wave, making it less likely to producea loss in the signal.

Among the respective embodiments described above, by combining thoseembodiments which can be combined, effects in the respective embodimentscan be synergistically combined. For example, the configuration foradding a bias current shown in the second, fourth, eighth, and ninthembodiments can be combined with a configuration having current mirrorcircuits at multiple stages shown in the fifth to seventh embodiments,and moreover, transconductance amplifiers provided thereby can of coursebe designed in a dual configuration, as shown in the second embodiment.These configurations are also included in the present invention.

Tenth Embodiment

FIG. 29 is a diagram showing a tenth embodiment. In this embodiment, thetransconductance amplifiers in the first to ninth embodiment are used infilter circuits. As shown in FIG. 29 a, in this embodiment, first-orderfilter 241 is connected in series with fourth-order filters 242, 243.

Each filter is a G_(m)-C filter which comprises a transconductanceamplifier, in any of the configurations of the first to ninthembodiments, and a capacitance. First-order filter 241 comprisestransconductance amplifiers 244, 245 and a capacitance as shown in FIG.29 b, while fourth-order filters 242, 243 each comprise fourtransconductance amplifiers 246-249 and a capacitance as shown in FIG.29 c.

Transconductance amplifier 244 which forms part of first-order filter241 has an output terminal and an inverting output terminal connected toan input terminal and an inverting input terminal of transconductanceamplifier 245, respectively, and grounded through capacitances as well.Transconductance amplifier 245 in turn has an output terminal and aninverting output terminal connected to the inverting input terminal andinput terminal of transconductance amplifier 245, and is thus appliedwith negative feedback.

Transconductance amplifier 246 which forms part of fourth-order filter242 or 243 has an output terminal and an inverting output terminalconnected to an input terminal and an inverting input terminal oftransconductance amplifier 248, respectively, and grounded throughcapacitances as well. Transconductance amplifier 248 has an outputterminal and an inverting output terminal connected to an input terminaland an inverting input terminal of transconductance amplifier 249,respectively, and grounded through capacitances as well.Transconductance amplifier 249 has an output terminal and an invertingoutput terminal connected to the inverting input terminal and inputterminal of transconductance amplifier 249, and is thus applied withnegative feedback. Transconductance amplifier 247 has an input terminaland an inverting input terminal connected to the output terminal andinverting output terminal of transconductance amplifier 248, andtransconductance amplifier 247 has an output terminal and an invertingoutput terminal connected to the inverting input terminal and inputterminal of transconductance amplifier 248.

By configuring first-order filter 241 and fourth-order filters 242, 243using the transconductance amplifiers of the first to ninth embodimentsand the capacitances, the filters can be configured with reducedcommon-mode components in signals which appear at their outputterminals. Also, it is not essential to combine first-order filter 241with fourth-order filters 242, 243, but they may be used as discretefilters, as a matter of course.

Eleventh Embodiment

FIG. 30 is a diagram showing an eleventh embodiment. In this embodiment,a Gm-C type current controlled oscillator is configured using thetransconductance amplifier of the ninth embodiment, and is applied to afrequency control loop to configure a PLL circuit.

FIG. 30 a shows the configuration of a PLL circuit which uses afrequency control loop. The PLL circuit of this embodiment comprisesphase detector 251, charge pump circuit 252, loop filter 253,voltage/current converter 254, current controlled oscillator 255, andcore filter 256.

Phase detector 251 receives reference frequency signal S1 from theoutside and current controlled oscillator 255 to generate a signal inaccordance with a phase difference therebetween. An output signal ofphase detector 251 is amplified by charge pump circuit 252, converted toa current in voltage/current converter 254 after high frequencycomponents thereof are removed by loop filter 253, and supplied tocurrent controlled oscillator 255 and core filter 256 as current controlsignal S2.

Current controlled oscillator 255 has its oscillation frequencycontrolled in accordance with the value of current control signal S2,while core filter 256 changes a frequency response characteristic inaccordance with the value of current control signal S2.

FIG. 30 b is a circuit diagram showing the configuration of currentcontrolled oscillator 255.

Current controlled oscillator 255 shown in FIG. 30 b comprisescomparison voltage generating circuit 257, comparators 258 ₁, 258 ₂, RSflip-flop 259, resistors R₂, R₁, R₂ provided between a power supply anda ground, and a switch, whose open/close state is controlled by RSflip-flop 259, for selectively supplying comparison voltage generatingcircuit 257 with a voltage divided by each resistor. The comparisonvoltage generating circuit is made up of transconductance amplifiergm_(m) and capacitance C_(m).

Comparators 258 ₁, 258 ₂, provided in front of RF flip-flop 259, compareoutput voltage V_(gm) of comparison voltage generating circuit 257 withV_(h) and V_(l), and switch the stage of RS flip-flop 259 in accordancewith the result. RS flip-flop 259 is set and reset to cause a change inits output, to change an input voltage to comparison voltage generatingcircuit 257, charging and discharging capacitance C_(m) to cause achange in output voltage V_(gm) of comparison voltage generating circuit257. This operation is repeated every half period of the oscillationfrequency of current controlled oscillator 255, and the output of RSflip-flop 259 is supplied to phase detector 251 as the output of currentcontrolled oscillator 255.

Oscillation frequency t_(OSC) of current controlled oscillator 255 isestimated by:

t _(OSC)=2×(1/(gm _(u) /C _(m))×(R ₁/(R ₁+2×R ₂))+t _(d))

where gm_(u) represents mutual conductance of transconductance amplifiergm_(m), and t _(d) represents a delay on a switching path indicated bythe broken arrow in FIG. 30 b. Oscillation frequency t_(OSC) of currentcontrolled oscillator 255 is dominated by the ratio gm_(u)/C_(m) of themutual transconductance to the capacitance, and by delay t_(d) on theswitching path. The ratio of the resistances ₁/(R₁+2×R₂) determines theratio of input and output voltages to comparison voltage generatingcircuit 257, and does not directly relate to oscillation frequencyt_(OSC) of current controlled oscillator 255. Stated another way,current controlled oscillator 255 will never be affected by processingsteps, temperature, or supplied voltage. This means that an adjustedoscillation accuracy is ideal.

FIG. 30 c is a circuit diagram specifically showing the configuration ofcomparison voltage generating circuit 257 shown in FIG. 30 b.

Comparison voltage generating circuit 257 is made up of transconductanceamplifiers 260, 261 which comprises transconductance amplifier gm_(m),and capacitance C_(m). Transconductance amplifier 260 has an outputterminal and an inverting output terminal connected to an input terminaland an inverting input terminal of transconductance amplifier 261, andalso connected to an inverting input terminal and an input terminal oftransconductance amplifier 260 and is thus applied with a negativefeedback. Transconductance amplifier 260 has the input terminal andinverting input terminal connected to an input terminal of comparisonvoltage generating circuit 257 through capacitances C1, C2,respectively, and an output terminal and an inverting output terminal oftransconductance amplifier 261 are respectively used as output terminalsof comparison voltage generating circuit 257.

Each of transconductance amplifiers 260, 261 is the transconductanceamplifier according to the ninth embodiment, and is supplied withcurrent control signal S2 to control signal input terminal s currentI_(ref). By connecting transconductance amplifier 260 to apply the samewith a negative feedback in the foregoing manner, a current flowingthrough an output stage is controlled in accordance with the value ofcurrent control signal S2 to control a signal bias at the output. As aresult of controlling the signal bias, the mutual conductance changes tocause a change in gm_(u)/C_(m) which is the ratio of the mutualconductance to the capacitance, dominating oscillation frequency t_(OSC)of current controlled oscillator 255, resulting in a change inoscillation frequency t_(OSC) of current controlled oscillator 255.

While this embodiment has been described on the assumption that thetransconductance amplifiers shown in the ninth embodiment are used astransconductance amplifiers 260, 261, the transconductance amplifiersshown in the second embodiment, fourth embodiment, and eighth embodimentcan be used as well. By connecting these transconductance amplifierslikewise so as to apply a negative feedback thereto, a current flowingthrough the output stage is controlled in accordance with voltage V_(c),causing a change in the mutual conductance. By converting currentcontrol signal S2 to a voltage through a resistor, and by supplyingvoltage V_(c), a signal bias generating circuit, a current controlledoscillator, and a PLL circuit can be configured in a manner similar tothis embodiment.

Further, the circuit characteristics are improved by usingtransconductance amplifiers which exhibit the same characteristics fortransconductance amplifiers 260, 261, but it is not essential to designthe transconductance amplifiers shown in the second, fourth, eighth, andninth embodiments. Transconductance amplifier 260 is provided for thepurpose of setting an input bias, and a similar circuit operation can beperformed, for example, by providing a capacitance between outputs oftransconductance amplifier 260. In configuring transconductanceamplifier gm_(m) in this embodiment, the importance to recognize thatthe transconductance amplifier, shown in the second, fourth, eighth,ninth embodiment, is used as transconductance amplifier 260 which isconnected so that it can be applied with a negative feedback to controlcurrent flowing through the output stage in accordance with the value ofcurrent control signal S2, such that the signal bias of the output iscontrolled.

Twelfth Embodiment

FIG. 31 is a circuit diagram showing the configuration of a twelfthembodiment according to the present invention.

This embodiment is made up of n-mos transistors M_(03A), M_(3A), M_(5A),M_(6A), M_(7A), M_(03B), M_(3B), M_(5B), M_(6B), M_(7B), and p-mostransistors M_(04A), M_(4A), M_(8A), M₀₄, M_(4B), M_(8B).

P-mos transistors M_(4A), M_(04A), M_(04B), M_(04B) correspond to n-mostransistors M_(3A), M_(03A), M_(03B), M_(3B), respectively, where thesecorresponding transistors are provided between a power supply and aground with their drains being in common. P-mos transistors M_(4A),M_(04A), M_(04B), M_(04B) have their gates connected to the drains ofp-mos transistors M_(04A), M_(04B) to configure a current mirrorcircuit. N-mos transistors M_(3A), M_(03A) have their gates connectedcommonly to gate of M_(03B), M_(3B) to configure an OTA similar to theOTA shown in FIG. 1 a.

Source-grounded n-mos transistors M_(5A), M_(6A), M_(5B), M_(6B)respectively form part of a current mirror circuit, where transistorsM_(5A), M_(6A) have their gates connected commonly to a drain oftransistor M_(6A), while transistors M_(5B), M_(6B) have their gatesconnected commonly to a drain of transistor M_(6B). Transistor M_(5A)has a drain connected to the drains of transistors M_(3A), M_(4A) whichserve as an output node of V_(OUT)+ of the OTA, while transistor M_(5B)has a drain connected to the drains of transistors M_(3B), M_(4B) whichserve as an output node of V_(out)− of the OTA.

Transistors M_(7A), M_(7B), M_(8A), M_(8B) form part of a CMFB circuit,where transistor M_(7A) supplied with reference signal V_(Y) at a gatehas a grounded source, and transistor M_(8A) having a gate used as nodeV_(X) (next stage) which is a feedback signal input terminal, has asource connected to the power supply. Transistors M_(7A), M_(8A) havetheir drains connected to the gates of n-mos transistors M_(5A), M_(6A)and to a drain of transistor M_(6A) to supply a reference current of acurrent mirror circuit which is made up of transistors M_(5A), M_(6A).

Transistor M_(7B) supplied with reference signal V_(Y) at a gate has agrounded source, and transistor M_(8B) having a gate used as node V_(X)(next stage), has a source connected to the power supply. TransistorsM_(7B), M_(8B) have their drains connected to the gates of n-mostransistors M_(5B), M_(6B) and to a drain of transistor M_(6B) to supplya reference signal to a current mirror circuit which is made up oftransistors M_(5B), M_(6B).

In the circuit configured as described above, the operation in an OTAportion is similar to the operation which has been described withreference to FIG. 1 a.

A description will be given of the operation of the circuit of thisembodiment which is connected as shown in FIG. 2 c to increasecommon-mode voltages of V_(IN)+, V_(IN)− supplied from the OTA at thepreceding stage.

As gate voltages of transistors M_(03A), M_(03B) increase due to anincrease in common-mode components of V_(IN)+, V_(IN)− supplied from theOTA at the preceding stage, drain currents of transistors M_(3A), M_(3B)increase, while voltage decreases at node V_(X) (preceding stage), whichis a feedback signal output terminal. Since node V_(X) (preceding stage)is connected to node V_(X) (next stage) of the OTA disposed at thepreceding stage, gate voltages decrease in transistor M_(8A), M_(8B) ofthe OTA at the preceding stage, while drain currents of transistorsM_(8A), M_(8B) increase. In this event, the difference between a currentwhich flows through transistors M_(7A), M_(7B), whose drain current isdetermined at a predetermined value by reference signal V_(Y), and acurrent which flows through transistors M_(8A), M_(8B), flows intotransistors M_(6A), M_(6B) to increase the current which flows intotransistors M_(6A), M_(6B). Since transistors M_(6A), M_(6B) form partof a current mirror circuit together with transistors M_(5A), M_(5B), acurrent flowing through transistors M_(5A), M_(5B) also increases to theaccompaniment of an increase in the current which flows throughtransistors M_(6A), M_(6B). As the current which flows throughtransistors M_(5A), M_(5B) increases, the voltages of V_(OUT)+, V_(OUT)−decrease. Since V_(OUT)+ and V_(OUT)− of the OTA at the preceding stageare V_(IN)+, V_(IN)− of the OTA at the next stage, a feedback circuit isconfigured.

In this embodiment configured as described above, the drains oftransistors M_(3A), M_(4A), M_(5A) or the drains of transistors M_(3B),M_(4B), M_(5B) are connected to each output node of the OTA, and threetransistors are connected to the output node. As a result, the connectedquantity is reduced from before to reduce output conductance andparasitic capacitance caused by each transistor connected in parallel,thus making it possible to prevent a reduction in output impedance ofthe OTA, and a degradation in characteristics as the OTA.

Thirteenth Embodiment

FIG. 32 is a circuit diagram showing the configuration of a thirteenthembodiment according to the present invention.

In this embodiment, n-mos transistor M9 supplied with reference signalV_(Y) at a gate is provided between each gate of p-mos transistorsM_(4A), M_(04A), M_(04B), M_(4B) and a ground in the circuit of thetwelfth embodiment shown in FIG. 31, to supply V_(IN)+, V_(IN)− to thesources of transistors M_(7A), M_(7B) which are supplied with referencesignal V_(Y) in the first embodiment.

In this embodiment configured as described above, an common-mode bias ofoutputs V_(OUT)+, V_(OUT)− is determined to a predetermined bias byn-mos transistor M9. Also, as common-mode components of V_(IN)+, V_(IN)−supplied from the OTA at the preceding stage increase to increase thegate voltages of transistors M_(03A), M_(03B), the drain currents oftransistors M_(03A), M_(03B) increase, while a voltage at node V_(X)(preceding stage) decreases. Since node V_(X) (preceding stage) isconnected to node V_(X) (next stage) of the OTA disposed at thepreceding stage, the gate voltages of transistors M_(8A), M_(8B)decrease in the OTA at the preceding stage, while the drain currents oftransistors M_(8A), M_(8B) increase. In this event, a current flowinginto transistors M_(6A), M_(6B) further increases due to an increase inthe gate voltage of transistors M_(7A), M_(7B) which are supplied withV_(IN)+, V_(IN)− at the gates, where the amount of current amounts toapproximately twice as much as the twelfth embodiment. Since transistorsM_(6A), M_(6B) form part of a current mirror circuit together withtransistors M_(5A), M_(5B), a current flowing through transistorsM_(5A), M_(5B) also increases to the accompaniment of an increase in thecurrent which flows through transistors M_(6A), M_(6B). As the currentflowing through transistors M_(5A), M_(5B) increases, the voltages ofV_(OUT)+ and V_(OUT)− decrease.

In this embodiment configured as described above, three transistors areconnected to each output node of the OTA, making it possible to preventa reduction in output impedance of the OTA and a degradation in thecharacteristics as the OTA, as is the case with the twelfth embodiment.Further, the signal amplitude of a response component in V_(OUT)+,V_(OUT)− to a change in common-mode components of V_(IN)+, V_(IN)−increases to approximately twice as much as the twelfth embodiment, thusimproving the feedback response speed.

Fourteenth Embodiment

FIG. 33 is a circuit diagram showing the configuration of a fourteenthembodiment according to the present invention.

In this embodiment, p-mos transistor M3′ supplied with reference signalV_(Y) at a gate is provided between each gate of p-mos transistorsM_(4A), M_(04A), M_(04B), M_(04B) and the power supply in the circuitshown in FIG. 2B, and transistors M_(3A)′, M_(3B)′ supplied withreference voltage V_(Y) are removed in the circuit shown in FIG. 2 b.

In this embodiment configured as described above, an common-mode bias ofoutputs V_(OUT)+, V_(OUT)− is set to a predetermined bias by p-mostransistor M3′. In this way, three transistors are connected to eachoutput node of the OTA, as is the case with the twelfth and thirteenthembodiments, making it possible to prevent a reduction in the outputimpedance of the OTA and a degradation in the characteristics as theOTA.

While each embodiment has been described on the assumption that thetransistors which make up circuits comprise p-mos transistors and n-mostransistors, the circuits may be made up of JFETs and bipolartransistors. Also, p-mos transistors may be replaced with n-mostransistors, and n-mos transistors may be replaced with p-mostransistors.

Alternatively, a well separation type, which is effective in reducingnoise, can be used for the n-mos transistors.

Fifteenth Embodiment

FIG. 34 is a diagram showing a fifteenth embodiment. In this embodiment,the transconductance amplifiers of the twelfth to fourteenth embodimentsare used in filter circuits. As shown in FIG. 34 a, this embodimentcomprises first-order filter 241 connected in series with fourth-orderfilters 242, 243.

Each filter is a G_(m)-C filter which comprises which comprises atransconductance amplifier, in any of the configurations of the twelfthto fourteenth embodiments, and a capacitance. First-order filter 241comprises transconductance amplifiers 244, 245 and a capacitance asshown in FIG. 34 b, while fourth-order filters 242, 243 each comprisefour transconductance amplifiers 246-249 and a capacitance as shown inFIG. 34 c.

Transconductance amplifier 244 which forms part of first-order filter241 has an output terminal and an inverting output terminal connected toan input terminal and an inverting input terminal of transconductanceamplifier 245, respectively, and is grounded through capacitances aswell. Transconductance amplifier 245 in turn has an output terminal andan inverting output terminal connected to the inverting input terminaland input terminal of transconductance amplifier 245, and is thusapplied with negative feedback. Also, V_(X) (preceding stage) oftransconductance amplifier 245 is connected to V_(X) (next stage) oftransconductance amplifier 244.

Transconductance amplifier 246 which forms part of fourth-order filter242 or 243 has an output terminal and an inverting output terminalconnected to an input terminal and an inverting input terminal oftransconductance amplifier 248, respectively, and is grounded throughcapacitances as well. Transconductance amplifier 248 has an outputterminal and an inverting output terminal connected to an input terminaland an inverting input terminal of transconductance amplifier 249,respectively, and is grounded through capacitances as well.Transconductance amplifier 249 has an output terminal and an invertingoutput terminal connected to the inverting input terminal and inputterminal of transconductance amplifier 249, and is thus applied withnegative feedback. Transconductance amplifier 247 has an input terminaland an inverting input terminal connected to the output terminal andinverting output terminal of transconductance amplifier 248, andtransconductance amplifier 247 has an output terminal and an invertingoutput terminal connected to the inverting input terminal and inputterminal of transconductance amplifier 248. Also, the V_(X) (precedingstage) of transconductance amplifier 248 is connected to the V_(X) (nextstage) of transconductance amplifiers 246, 247, and the V_(X) (nextstage) of transconductance amplifier 248 is connected to the V_(X)(preceding stage) of transconductance amplifiers 248, 249.

By configuring first-order filter 241 and fourth-order filters 242, 243using the transconductance amplifiers of the first to third embodimentsand capacitances, the filters can be configured with the common-modebias being set at a predetermined value such that there is limiteddegradation in characteristics. Also, it is not essential to combinefirst-order filter 241 with fourth-order filters 242, 243, but they maybe used as discrete filters, as a matter of course.

Sixteenth Embodiment

FIG. 35 is a diagram showing a sixteenth embodiment. In this embodiment,a Gm-C type current controlled oscillator is configured using thetransconductance amplifier of the twelfth to fourteenth embodiments, andis applied to a frequency control loop to configure a PLL circuit.

FIG. 35 a shows the configuration of a PLL circuit which uses afrequency control loop. The PLL circuit of this embodiment comprisesphase detector 251, charge pump circuit 252, loop filter 253,voltage/current converter 254, current controlled oscillator 255, andcore filter 256.

Phase detector 251 receives reference frequency signal S1 from theoutside and current controlled oscillator 255 to generate a signal inaccordance with a phase difference therebetween. The output signal ofphase detector 251 is amplified by charge pump circuit 252, converted toa current in voltage/current converter 254 after high frequencycomponents thereof are removed by loop filter 253, and is supplied tocurrent controlled oscillator 255 and core filter 256 as current controlsignal S2.

Current controlled oscillator 255 has its oscillation frequencycontrolled in accordance with the value of current control signal S2,while core filter 256 changes the frequency response characteristics inaccordance with the value of current control signal S2.

FIG. 35 b is a circuit diagram showing the configuration of currentcontrolled oscillator 255.

Current controlled oscillator 255 shown in FIG. 35 b comprisescomparison voltage generating circuit 257, comparators 258 ₁, 258 ₂, RSflip-flop 259, resistors R₂, R₁, R₂ provided between a power supply anda ground, and a switch whose open/close state of which is controlled byRS flip-flop 259, for selectively supplying comparison voltagegenerating circuit 257 with a voltage divided by each resistor.Comparison voltage generating circuit 257 is made up of transconductanceamplifier gm_(m) and capacitance C_(m).

Comparators 258 ₁, 258 ₂, provided in front of RF flip-flop 259, compareoutput voltage V_(gm) of comparison voltage generating circuit 257 withV_(h) and V_(l), and switch the stage of RS flip-flop 259 in accordancewith the result. RS flip-flop 259 is set and reset to cause a change inits output, to change an input voltage to signal generating bias circuit257, charging and discharging capacitance C_(m) to cause a change inoutput voltage V_(gm) of comparison voltage generating circuit 257. Thisoperation is repeated every half period of the oscillation frequency ofcurrent controlled oscillator 255, and the output of RS flip-flop 259 issupplied to phase detector 251 as the output of current controlledoscillator 255.

Oscillation frequency t_(OSC) of current controlled oscillator 255 isestimated by:

t _(OSC)=2×(1/(gm _(u) /C _(m))×(R ₁/(R ₁+2×R ₂))+t _(d))

where gm_(u) represents mutual conductance of transconductance amplifiergm_(m), and t _(d) represents a delay on a switching path indicated bythe broken arrow in FIG. 35 b. Oscillation frequency t_(OSC) of currentcontrolled oscillator 255 is dominated by the ratio gm_(u)/C_(m) of themutual transconductance to the capacitance, and by delay t_(d) on theswitching path. The ratio of the resistances R₁/(R₁+2×R₂) determines theratio of input and output voltages to comparison voltage generatingcircuit 257, and does not directly relate to oscillation frequencyt_(OSC) of current controlled oscillator 255. Stated another way,current controlled oscillator 255 will never be affected by processingsteps, temperature, or supplied voltage. This means that an adjustedoscillation accuracy is ideal.

In this embodiment, the transconductance amplifier of the first to thirdembodiments is used for comparison voltage generating circuit 257, andcurrent control signal S2 is converted to a voltage by a resistor (notshown) and supplied to V_(X) (next stage) which is a feedback signalinput terminal. By applying a negative feedback as described above, acurrent flowing through the output stage is controlled in accordancewith the value of current control signal S2 to control the common-modebias of the output. As a result of controlling the common-mode bias, themutual conductance changes to cause a change in gm_(u)/C_(m) which isthe ratio of the mutual conductance to the capacitance, dominatingoscillation frequency t_(OSC) of current controlled oscillator 255,resulting in a change in oscillation frequency t_(OSC) of currentcontrolled oscillator 255.

1-30. (canceled)
 31. A voltage-to-current converting method forgenerating a first current and a second current proportional to adifference between an input first voltage signal and second voltagesignal, characterized by comprising the steps of: converting the firstvoltage signal to a first current signal; converting the second voltagesignal to a second current signal; generating an common-mode componentof the first current signal and the second current signal; generating athird current signal and a fourth current signal by subtracting thecommon-mode component from the first current signal and the secondcurrent signal, respectively, generating a first output by subtractingthe fourth current signal from the third current signal, and generatinga second output by subtracting the third current signal from the fourthcurrent signal.
 32. A transconductance amplifier characterized bycomprising: a first and a second voltage/current converting element forconverting a first voltage signal to a current signal; a third and afourth voltage/current converting element for converting a secondvoltage signal to a current signal; an common-mode current generatingpart for converting each first voltage signals and each second voltagesignal to a current signal, and further generating an common-modecurrent in accordance with an common-mode component of each currentsignal; a first current circuit for subtracting the common-modecomponent by using said common-mode component generating part from eachcurrent signal converted by each of said first to fourth voltage/currentconverting elements; a second current circuit for supplying a differencebetween a current signal by using said first voltage/current convertingelement from which the common-mode component is subtracted by using saidfirst current circuit and a current signal by using said thirdvoltage/current converting element as a first current output; and athird current circuit for supplying a difference between a currentsignal by using said fourth voltage/current converting element fromwhich the common-mode component is subtracted by using said firstcurrent circuit and a current signal by using said secondvoltage/current converting element as a second current output.
 33. Thetransconductance amplifier according to claim 32, characterized in thatsaid common-mode current generating part comprises a fifthvoltage/current converting element and a sixth voltage/currentconverting element for converting the first voltage signal and thesecond voltage signal to current signals, respectively.
 34. Thetransconductance amplifier according to claim 33, characterized in that:said first to sixth voltage/current converting elements comprises afirst to a sixth first conductivity type transistor supplied with thefirst voltage signal or the second voltage signal at bases or gates,said first current circuit comprises a plurality of second conductivitytype transistors, said plurality of second conductivity type transistorshave their gates in common, and at least one of said plurality of secondconductivity type transistors have a gate short-circuited to a drain,and an output of said second conductivity type transistor is connectedto any of the outputs of said first to sixth voltage/current convertingelements.
 35. The transconductance amplifier according to claim 33,characterized in that: said first to sixth voltage/current convertingelements comprise a first to a sixth first conductivity type transistorsupplied with the first voltage signal or the second voltage signal atbases or gates, said first current circuit comprises a first to a sixthsecond conductivity type transistor provided between a power supply anda ground together with said first to sixth voltage/current convertingelements, said second conductivity type transistors have their gates andsources in common, and at least one of said second conductivity typetransistors has a gate short-circuited to a drain, and outputs of saidsecond conductivity type transistor are connected to outputs of saidfirst to sixth voltage/current converting elements, respectively. 36.The transconductance amplifier according to claim 35, characterized inthat: said fifth first conductivity type transistor and said sixth firstconductivity type transistor are first transistors which have the samesize with each other, said first to fourth first conductivity typetransistors are second transistors which have the same size with oneanother, said fifth second conductivity type transistor and said sixthsecond conductivity type transistor are third transistors which have thesame size with each other, said first to fourth second conductivity typetransistors are fourth transistors which have the same size with oneanother, and the ratio of the size of said first transistor to the sizeof said second transistor is equal to the ratio of the size of saidthird transistor to the size of said fourth transistor.
 37. Thetransconductance amplifier according to claim 35, characterized in that:said first first conductivity type transistor forms part of a firstcurrent output supplying part, and said fourth first conductivity typetransistor forms part of a second current output supplying part, saidsecond current circuit comprises a seventh first conductivity typetransistor having an output in common with an output of said first firstconductivity type transistor, and an eighth first conductivity typetransistor having an output and a gate in common with an output of saidthird first conductivity type transistor and a gate of said seventhfirst conductivity type transistor, said third current circuit comprisesa ninth first conductivity type transistor having an output in commonwith an output of said second first conductivity type transistor, and atenth first conductivity type transistor having an output and a gate incommon with an output of said fourth first conductivity type transistorand a gate of said ninth first conductivity type transistor, saidsecond, third, fifth, and sixth first conductivity type transistors arefirst transistors which match in size with one another, said first firstconductivity type transistor and said fourth first conductivity typetransistor are second transistors which have the same size with eachother, said eighth first conductivity type transistor and said tenthfirst conductivity type transistor are third transistors which have thesame size with each other, said seventh first conductivity typetransistor and said ninth first conductivity type transistor are fourthtransistors which have the same size with each other, said fifth secondconductivity type transistor, said sixth second conductivity typetransistor, said second second conductivity type transistor, and saidthird second conductivity type transistor are fifth transistors whichhave the same size with one another, said first second conductivity typetransistor and said fourth second conductivity type transistor are sixthtransistors which have the same size with each other, and the ratio ofthe size of said first transistor to the size of said second transistor,the ratio of the size of said third transistor to the size of saidfourth transistor, and the ratio of the size of said fifth transistor tothe size of said sixth transistor are equal.
 38. The transconductanceamplifier according to claim 33, characterized in that: said firstcurrent circuit comprises a plurality of sets of said first to sixthsecond conductivity type transistors, wherein second conductivity typetransistors in other sets are configured such that outputs of saidsecond conductivity type transistors are connected to inputs of saidsecond conductive type transistors.
 39. The transconductance amplifieraccording to claim 33, characterized in that: said transconductanceamplifier comprises a plurality of sets of the first to fourth firstconductivity type transistors making up said second current circuit andthird current circuits, wherein the first conductivity type transistorsin other sets are connected to outputs of said second conductivity typetransistors and to inputs of said second conductivity type transistors.40. A transconductance amplifier characterized by comprising: aplurality of transconductance amplifiers according to claim 32; a fourthcurrent circuit provided in one transconductance amplifier forgenerating a difference between a first current output and a secondcurrent output in said one transconductance amplifier as a first currentoutput; and a fifth current circuit provided in another transconductanceamplifier for generating a difference between a second current outputand a first current output in the other transconductance amplifier as asecond current output.
 41. The transconductance amplifier according toclaim 33, characterized in that: said common-mode current generatingpart comprises a seventh voltage/current converting element suppliedwith a third voltage signal at a base or a gate, and generates ancommon-mode current including a bias current in accordance with thethird voltage signal as the common-mode current.
 42. Thetransconductance amplifier according to claim 33, characterized bycomprising: a first bias current generating element for generating afirst bias current added to a reference current supplied by said firstcurrent circuit to said second current circuit; and a second biascurrent generating element for generating a second bias current added toa reference current supplied by said first current circuit to said thirdcurrent circuit.
 43. A first-order filter circuit configured using thetransconductance amplifiers according to claim 32, wherein: said filtercircuit comprises said transconductance amplifier and a capacitance, andis made up of a first and a second transconductance amplifier, saidfirst transconductance amplifier having an output terminal and aninverting output terminal connected to an input terminal and aninverting input terminal of said second transconductance amplifier,respectively, and grounded through capacitances, said secondtransconductance amplifier having an output terminal and an invertingoutput terminal connected to the inverting input terminal and inputterminal of said second transconductance amplifier.
 44. A fourth-orderfilter circuit configured using the transconductance amplifiersaccording to claim 32, wherein: said filter circuit comprises a first toa fourth transconductance amplifier, said first transconductanceamplifier having an output terminal and an inverting output terminalconnected to an input terminal and an inverting input terminal of saidsecond transconductance amplifier and grounded through capacitances,said second transconductance amplifier having an output terminal and aninverting output terminal connected to the input terminal and invertinginput terminal of said second transconductance amplifier and groundedthrough capacitances, said third transconductance amplifier having anoutput terminal and an inverting output terminal connected to aninverting input terminal and an input terminal of said thirdtransconductance amplifier, said fourth transconductance amplifierhaving an input terminal and an inverting input terminal connected tothe output terminal and inverting output terminal of said secondtransconductance amplifier, said fourth transconductance amplifierhaving an output terminal and an inverting output terminal connected tothe inverting input terminal and input terminal of said secondtransconductance amplifier.
 45. A filter circuit comprising: afirst-order filter circuit comprising first and second transconductanceamplifiers, said first transconductance amplifier having an outputterminal and an inverting output terminal connected to an input terminaland an inverting input terminal of said second transconductanceamplifier, respectively, and grounded through capacitances, said secondtransconductance amplifier having an output terminal and an invertingoutput terminal connected to the inverting input terminal and inputterminal of said second transconductance amplifier; and two fourth-orderfilter circuits connected in series with said first-order filtercircuit, each of said fourth-order filter circuit comprising third tosixth transconductance amplifiers, wherein: said third transconductanceamplifier having an output terminal and an inverting output terminalconnected to an input terminal and an inverting input terminal of saidfourth transconductance amplifier and grounded through capacitances,said fourth transconductance amplifier having an output terminal and aninverting output terminal connected to the input terminal and invertinginput terminal of said fourth transconductance amplifier and groundedthrough capacitances, said fifth transconductance amplifier having anoutput terminal and an inverting output terminal connected to aninverting input terminal and an input terminal of said fifthtransconductance amplifier, said sixth transconductance amplifier havingan input terminal and an inverting input terminal connected to theoutput terminal and inverting output terminal of said fourthtransconductance amplifier, said sixth transconductance amplifier havingan output terminal and an inverting output terminal connected to theinverting input terminal and input terminal of said fourthtransconductance amplifier, wherein each of said transconductanceamplifiers comprises, a first and a second voltage/current convertingelement for converting a first voltage signal to a current signal; athird and a fourth voltage/current converting element for converting asecond voltage signal to a current signal; an common-mode currentgenerating part for converting each first voltage signals and eachsecond voltage signal to a current signal, and further generating ancommon-mode current in accordance with an common-mode component of eachcurrent signal; a first current circuit for subtracting the common-modecomponent by using said common-mode component generating part from eachcurrent signal converted by each of said first to fourth voltage/currentconverting elements; a second current circuit for supplying a differencebetween a current signal by using said first voltage/current convertingelement from which the common-mode component is subtracted by using saidfirst current circuit and a current signal by using said thirdvoltage/current converting element as a first current output; and athird current circuit for supplying a difference between a currentsignal by using said fourth voltage/current converting element fromwhich the common-mode component is subtracted by using said firstcurrent circuit and a current signal by using said secondvoltage/current converting element as a second current output.
 46. Avoltage generating circuit configured using the transconductanceamplifier according to claim 41, characterized in that: saidtransconductance amplifier has an output terminal and an invertingoutput terminal connected to an inverting input terminal and an inputterminal, and said voltage generating circuit comprises a capacitancefor alternatingly grounding one output part of said transconductanceamplifier.
 47. A voltage generating circuit configured using thetransconductance amplifier according to claim 41, characterized in that:said voltage generating circuit comprises a first and a secondtransconductance amplifier and a capacitance, said firsttransconductance amplifier having an output terminal and an invertingoutput terminal connected to an input terminal and an inverting inputterminal of said second transconductance amplifier and connected to aninverting input terminal and an input terminal of said firsttransconductance amplifier, said input terminal and inverting inputterminal of said first transconductance amplifier connected to an inputthrough capacitances, respectively, an output terminal and an invertingoutput terminal of said second transconductance amplifier respectivelyserving as outputs.
 48. A current controlled oscillator configured usingthe voltage generating circuit according to claim 46, comprising: aplurality of resistors provided in series between a power supply and aground; a switch group provided between said plurality of resistors andan input of said voltage generating circuit for selectively applying avoltage divided by said plurality of resistors to said voltagegenerating circuit; a first and a second comparator for comparingterminal voltages of said plurality of resistors provided in series withan output of said voltage generating circuit; and a flip-flop whosestate changes in state in accordance with outputs of said first andsecond comparators and generates an output which defines an oscillationfrequency and is used as a switching control signal for said switchgroup.
 49. A PLL circuit configured using the current controlledoscillator according to claim 48, comprising: a current controlledoscillator, whose oscillation frequency is controlled by a currentcontrol signal; a phase detector for receiving a reference frequencysignal and an output of said current controlled oscillator to generate asignal in accordance with a phase difference therebetween; and avoltage/current converter for converting an output of said phasedetector to a current, and supplying the same to said current controlledoscillator.
 50. A transconductance amplifier for generating a firstoutput voltage signal and a second output voltage signal proportional toa difference between a first input voltage signal and a second inputvoltage signal applied thereto from a first and a second output stage,respectively, said transconductance amplifier characterized bycomprising: a feedback signal output terminal for outputting common-modecomponents of the first output voltage signal and second output voltagesignal; a feedback signal input terminal; and a reference signal inputterminal, provided in said first and second output stages, respectively;and feedback signal conveying means for controlling the first outputvoltage signal or second output voltage signal in accordance with aninput signal to said feedback signal input terminal and an input signalto said reference signal input terminal, wherein said feedback signalconveying means is connected to each output stage.
 51. Atransconductance amplifier for generating a first output voltage signaland a second output voltage signal proportional to a difference betweena first input voltage signal and a second input voltage signal appliedthereto from a first and a second output stage, respectively, saidtransconductance amplifier characterized by comprising: a feedbacksignal output terminal for outputting common-mode components of thefirst output voltage signal and second output voltage signal; areference signal input terminal for receiving a reference signal forbringing each of said first and second output stages into apredetermined bias state; a feedback signal input terminal; and feedbacksignal communicating means for controlling the first output voltagesignal or the second output voltage signal in accordance with an inputsignal to said feedback signal input terminal and an input signal tosaid reference signal input terminal, provided in said first and secondoutput stages, respectively; and feedback signal conveying means forcontrolling the first output voltage signal or second output voltagesignal in accordance with an input signal to said feedback signal inputterminal and an input signal to said reference signal input terminal,wherein said feedback signal conveying means are connected to eachoutput stage.
 52. The transconductance amplifier according to claim 50,wherein said feedback signal conveying means comprises: a current mirrorcircuit having an output part connected to said output stage; a firsttransistor of a first conductivity type having a control terminalconnected to said feedback signal input terminal; and a secondtransistor of a second conductivity type having a control terminalconnected to said reference signal input terminal for determining areference current for said current mirror circuit together with saidfirst transistor.
 53. The transconductance amplifier according to claim51, wherein said feedback signal conveying means comprises: a currentmirror circuit having an output part connected to said output stage; afirst transistor of a first conductivity type having a control terminalconnected to said feedback signal input terminal; and a secondtransistor of a second conductivity type for receiving the first inputvoltage signal or the first input voltage signal at a control terminalto determine a reference current for said current mirror circuittogether with said first transistor.
 54. The transconductance amplifieraccording to claim 51, wherein: said feedback signal conveying means isa transistor which has a control terminal connected to said feedbacksignal input terminal and an output part connected to said output stage.55. A first-order filter circuit configured using the transconductanceamplifier according to any one of claims 50 to 54, wherein: said filtercircuit comprises said transconductance amplifier and a capacitance, andis made up of a first and a second transconductance amplifier, saidfirst transconductance amplifier having an output terminal and aninverting output terminal connected to an input terminal and aninverting input terminal of said second transconductance amplifier,respectively, and grounded through capacitances, said secondtransconductance amplifier having an output terminal and an invertingoutput terminal connected to the inverting input terminal and inputterminal of said second transconductance amplifier.
 56. A fourth-orderfilter circuit configured using the transconductance amplifiersaccording to claim 50, wherein: said filter circuit comprises a first toa fourth transconductance amplifier, said first transconductanceamplifier having an output terminal and an inverting output terminalconnected to an input terminal and an inverting input terminal of saidsecond transconductance amplifier and grounded through capacitances,said second transconductance amplifier having an output terminal and aninverting output terminal connected to the input terminal and invertinginput terminal of said second transconductance amplifier and groundedthrough capacitances, said third transconductance amplifier having anoutput terminal and an inverting output terminal connected to aninverting input terminal and an input terminal of said thirdtransconductance amplifier, said fourth transconductance amplifierhaving an input terminal and an inverting input terminal connected tothe output terminal and inverting output terminal of said secondtransconductance amplifier, said fourth transconductance amplifierhaving an output terminal and an inverting output terminal connected tothe inverting input terminal and input terminal of said secondtransconductance amplifier.
 57. A filter circuit comprising: afirst-order filter circuit comprising first and second transconductanceamplifiers, wherein: said first transconductance amplifier having anoutput terminal and an inverting output terminal connected to an inputterminal and an inverting input terminal of said second transconductanceamplifier, respectively, and grounded through capacitances, said secondtransconductance amplifier having an output terminal and an invertingoutput terminal connected to the inverting input terminal and inputterminal of said second transconductance amplifier, and two fourth orderfilter circuits connected in series to said first-order filter circuit,each of said fourth-order filter circuits comprising third to sixthtransconductance amplifiers said third transconductance amplifier havingan output terminal and an inverting output terminal connected to aninput terminal and an inverting input terminal of said fourthtransconductance amplifier and grounded through capacitances, saidfourth transconductance amplifier having an output terminal and aninverting output terminal connected to the input terminal and invertinginput terminal of said fourth transconductance amplifier and groundedthrough capacitances, said fifth transconductance amplifier having anoutput terminal and an inverting output terminal connected to aninverting input terminal and an input terminal of said fifthtransconductance amplifier, said sixth transconductance amplifier havingan input terminal and an inverting input terminal connected to theoutput terminal and inverting output terminal of said fourthtransconductance amplifier, said sixth transconductance amplifier havingan output terminal and an inverting output terminal connected to theinverting input terminal and input terminal of said fourthtransconductance amplifier, wherein each of said transconductanceamplifiers comprises a transconductance amplifier according to any oneof claims 50-54.
 58. A voltage generating circuit configured using thetransconductance amplifier according to claim 50, characterized in that:a feedback signal input terminal is used as a control signal inputterminal for generating a bias current to change mutual conductance, andsaid voltage generating circuit comprises a capacitance foralternatingly grounding an output current.
 59. A current controlledoscillator configured using the voltage generating circuit according toclaim 58, comprising: a plurality of resistors provided in seriesbetween a power supply and a ground; a switch group provided betweensaid plurality of resistors and an input of said voltage generatingcircuit for selectively applying a voltage divided by said plurality ofresistors to said voltage generating circuit; a first and a secondcomparator for comparing terminal voltages of said plurality ofresistors provided in series with an output of said voltage generatingcircuit; and a flip-flop whose state changes in state in accordance withoutputs of said first and second comparators and generates an outputwhich defines an oscillation frequency and is used as a switchingcontrol signal for said switch group.
 60. A PLL circuit configured usingthe current controlled oscillator according to claim 58, comprising: acurrent controlled oscillator, whose oscillation frequency is controlledby a current control signal; a phase detector for receiving a referencefrequency signal and an output of said current controlled oscillator togenerate a signal in accordance with a phase difference therebetween;and a voltage/current converter for converting an output of said phasedetector to a current, and supplying the same to the control signalinput terminal of said current controlled oscillator.